Communication protocol in directional drilling system, apparatus and method utilizing multi-bit data symbol transmission

ABSTRACT

A system includes a transmitter for use in conjunction with a horizontal directional drilling system that transmits a multi-bit symbol stream that characterizes sensor symbols for receipt by an aboveground portable device. The portable device receives the symbol stream for aboveground recovery of the sensor signals. The transmitter can precisely place the symbol frequencies at least to avoid a noise environment, as well as to avoid powerline harmonics, and can utilize wave shaping for transmitted symbols at least to provide for transmission power control, spectral content control and wideband antenna matching. The receiver can measure the noise environment to identify the symbol frequencies used by the transmitter. The noise can be scanned at an incremental resolution across a wide frequency bandwidth for display or automatic symbol frequency selection.

RELATED APPLICATIONS

This application is a continuation application of copending U.S. patentapplication Ser. No. 14/845,231 filed on Sep. 3, 2015, which claimspriority from U.S. Provisional Patent Application Ser. No. 62/046,772,filed on Sep. 5, 2014, U.S. Provisional Patent Application Ser. No.62/080,225 filed on Nov. 14, 2014, and U.S. Provisional PatentApplication Ser. No. 62/143,104, filed on Mar. 4, 2015, each of which ishereby incorporated by reference in its entirety.

BACKGROUND

The present application is generally related to the field ofcommunications relating to an inground device and, more particularly, toadvanced inground device communication protocol using multi-bit datasymbol and associated methods.

A technique that is often referred to as horizontal directional drilling(HDD) can be used for purposes of installing a utility without the needto dig a trench. A typical utility installation involves the use of adrill rig having a drill string that supports a boring tool at a distalor inground end of the drill string. The drill rig forces the boringtool through the ground by applying a thrust force to the drill string.The boring tool is steered during the extension of the drill string toform a pilot bore. Upon completion of the pilot bore, the distal end ofthe drill string is attached to a pullback apparatus which is, in turn,attached to a leading end of the utility. The pullback apparatus andutility are then pulled through the pilot bore via retraction of thedrill string to complete the installation. In some cases, the pullbackapparatus can comprise a back reaming tool which serves to expand thediameter of the pilot bore ahead of the utility so that the installedutility can be of a greater diameter than the original diameter of thepilot bore.

Steering of a boring tool can be accomplished in a well-known manner byorienting an asymmetric face of the boring tool for deflection in adesired direction in the ground responsive to forward movement. In orderto control this steering, it is desirable to monitor the orientation ofthe boring tool based on sensor readings obtained by sensors that formpart of an electronics package that is supported by the boring tool. Thesensor readings, for example, can be modulated onto a locating signalthat is transmitted by the electronics package for reception aboveground by a portable locator or other suitable above ground device. Insome systems, the electronics package can couple a carrier signalmodulated by the sensor readings onto the drill string to then transmitthe signal to the drill rig by using the drill string as an electricalconductor. Irrespective of the manner of transmission of the sensor dataand for a given amount of transmission power, there is a limitedtransmission range at which the sensor data can be recovered withsufficient accuracy. The transmission range can be still further limitedby factors such as, for example, electromagnetic interference that ispresent in the operational region. One prior art approach, in attemptingto increase transmission range, is to transmit data from the boring toolor other inground tool at what Applicants refer to herein as a “magicfrequency.” The latter can be characterized as a carrier frequency thatremarkably avoids environmental interference to provide for effectivereception range despite the wide variety of environmental interferencethat may be encountered. As will be further discussed, Applicants submitthat such a magic frequency does not exist at least on the basis ofapplication to any sort of broad geographic region and particularly onthe basis of worldwide application. Another prior art approach is simplyto increase the transmission power. Applicants recognize, however, thatthis approach can be of limited value, particularly when the ingroundelectronics package is powered by batteries. Still another prior artapproach resides in lowering the data or baud rate at which data ismodulated onto the locating signal. Unfortunately, this approach isattended by a drop in data throughput.

The foregoing examples of the related art and limitations relatedtherewith are intended to be illustrative and not exclusive. Otherlimitations of the related art will become apparent to those of skill inthe art upon a reading of the specification and a study of the drawings.

SUMMARY

The following embodiments and aspects thereof are described andillustrated in conjunction with systems, tools and methods which aremeant to be exemplary and illustrative, not limiting in scope. Invarious embodiments, one or more of the above-described problems havebeen reduced or eliminated, while other embodiments are directed toother improvements.

In one aspect of the disclosure, a transmitter and associated method aredescribed for use in conjunction with a horizontal directional drillingsystem that includes a drill string that extends from a drill rig to aninground tool that supports the transmitter such that extension andretraction of the drill string moves the inground tool through theground during an inground operation. The transmitter includes an antennaand one or more sensors for generating one or more sensor signals. Aprocessor is configured for generating a multi-bit symbol stream basedon the sensor signals. An antenna driver arrangement is configured forelectrically driving the antenna to emit a dipole locating signal, as adepth signal, for aboveground reception at least for use in determininga depth of the inground tool and for electrically driving the antennabased on the multi-bit symbol stream to emit an electromagnetic symbolstream for aboveground recovery of the sensor signals.

In another aspect of the disclosure, a transmitter and associated methodare described for use in conjunction with a horizontal directionaldrilling system that includes a drill string that extends from a drillrig to an inground tool that supports the transmitter such thatextension and retraction of the drill string moves the inground toolthrough the ground during an inground operation. The transmitterincludes an antenna and one or more sensors for generating one or moresensor signals. A direct digital synthesizer is configured forgenerating a symbol stream, made up of a plurality of fixed-frequencysymbols, and the direct digital synthesizer is configured to customize adrive waveform shape for different symbol frequencies. An antenna driveris configured for electrically driving the antenna based on the symbolstream to emit an electromagnetic symbol stream for aboveground recoveryof the sensor signals.

In still another aspect of the disclosure, a transmitter and associatedmethod are described for use in conjunction with a horizontaldirectional drilling system that includes a drill string that extendsfrom a drill rig to an inground tool that supports the transmitter suchthat extension and retraction of the drill string moves the ingroundtool through the ground during an inground operation. The transmitterincludes an antenna and one or more sensors for generating one or moresensor signals. A processor is configured for generating a multi-bitsymbol stream based on the sensor signals. An antenna driver arrangementis configured for electrically driving the antenna based on themulti-bit symbol stream to emit an electromagnetic symbol stream atleast for aboveground recovery of the sensor signals.

In yet another aspect of the disclosure, a portable device andassociated method are described for use in conjunction with atransmitter that is configured to move through the ground in a regionduring an operational procedure while transmitting a transmitter signalthat is receivable by the portable device subject to electromagneticnoise that can vary within the region. The portable device includes areceiver configured to receive the transmitter signal as a multibitsymbol stream which at least characterizes a set of sensor informationrelating to the operation of the transmitter during the ingroundoperation to recover the set of sensor information.

In a continuing aspect of the disclosure, a portable device andassociated method are described for use in conjunction with atransmitter that is configured to move through the ground in a regionduring an operational procedure while transmitting a transmitter signalthat is receivable by the portable device subject to electromagneticnoise that can vary within the region. The portable device includes areceiver configured to (i) measure the electromagnetic noise andidentify a set of symbol frequencies in response to the measuredelectromagnetic noise for subsequent transmission from the transmitterto form a multi-bit symbol stream based on the set of symbolfrequencies, each of which multi-bit symbols corresponds to one of thesymbol frequencies, at least to characterize sensor information relatingto the operation of the transmitter, and (ii) receive the multibitsymbol stream from the transmitter during the inground operation torecover the sensor information.

In a further aspect of the disclosure, a portable device and associatedmethod are described for use in conjunction with a transmitter that isconfigured to move through the ground in a region during an operationalprocedure while transmitting a transmitter signal that is receivable bythe portable device subject to electromagnetic noise that can varywithin the region. The portable device includes a receiver configured to(i) measure the electromagnetic noise and identify a set of symbolfrequencies in response to the measured electromagnetic noise forsubsequent transmission from the transmitter at least to characterizesensor information relating to the operation of the transmitter, (ii)receive the symbol frequencies from the transmitter during the ingroundoperation to recover the sensor information and (iii) allocate atransmit power to each of the symbol frequencies.

In another aspect of the disclosure, a portable device and associatedmethod are described for use in conjunction with a transmitter that isconfigured to move through the ground in a region during an operationalprocedure while transmitting a transmitter signal that is receivable bythe portable device subject to electromagnetic noise that can varywithin the region. The portable device includes a receiver configured toreceive a packet structure from the transmitter that is made up of aplurality of multi-bit symbols in the symbol stream including at leastone group of multi-bit symbols characterizing the sensor information aspacket data and at least another group of multi-bit symbols serving as ablock of error correction data, the packet data characterizing a set ofsensor information relating to the operation of the transmitter duringthe inground operation. A slicer is configured for receiving eachmulti-bit symbol as a plurality of symbol slices that are spaced apartin time and each symbol slice includes a set of symbol frequencymagnitudes including a magnitude for each symbol frequency. Aspectrogram buffer includes a length made up of a series of slicepositions, each of which slice positions is configured to store one setof the symbol frequency magnitudes, and the length of the spectrogrambuffer is sufficient to store a total number of symbol slicescorresponding to a time duration of the packet structure. A time sliceswitch is configured for routing the symbol slices to the slicepositions of the spectrogram buffer to sequentially and selectivelystore the set of symbol frequency magnitudes associated with eachsuccessive symbol slice. A decoder is configured to detect, as part ofrecovering the packet data, a start symbol of the packet structure inthe spectrogram buffer based on the block of error correction data.

In yet another aspect of the disclosure, a system and associated methodare described for use with a horizontal directional drilling system thatincludes a drill string that extends from a drill rig to an ingroundtool such that extension and retraction of the drill string moves theinground tool through the ground during an inground operation. Thesystem includes a transmitter that is supported by the inground toolincluding an antenna, one or more sensors for generating one or moresensor signals, a processor configured for generating a multi-bit symbolstream based on the sensor signals, and an antenna driver forelectrically driving the antenna to emit an electromagnetic symbolstream based on the multi-bit symbol stream. The system further includesa portable device including a receiver configured to receive themultibit symbol stream in a normal mode during the inground operation torecover the set of sensor information subject to the electromagneticnoise.

In still another aspect of the disclosure, a transmitter and associatedmethod are described for use with a horizontal directional drillingsystem that includes a drill string that extends from a drill rig to aninground tool that supports the transmitter such that extension andretraction of the drill string moves the inground tool through theground during an inground operation. The transmitter includes an antennaand one or more sensors for generating one or more sensor signals. Amodulator is configured for generating at least one modulated datafrequency at a carrier frequency based on the sensor signals. A depthtone generator is configured for producing an unmodulated depth tonefrequency that is twenty or more times less than the carrier frequencyand an antenna driver for electrically driving at least one antenna toemit the depth tone frequency and the carrier frequency for abovegrounddetection of the depth tone frequency and for recovery of the sensorsignals from the modulated data frequency.

In a further aspect of the disclosure, a portable device and associatedmethod are described for use in conjunction with a horizontaldirectional drilling system that includes a drill string that extendsfrom a drill rig to an inground tool such that extension and retractionof the drill string moves the inground tool through the ground during aninground operation and the inground tool supports a transmitter thattransmits an electromagnetic signal for detection by the portable devicesuch that extension and retraction of the drill string moves theinground tool through the ground during an inground operation. Theportable device includes a receiver for detecting the signal with thetransmitter above ground in a first operational mode and below a surfaceof the ground in a second operational mode and a processor configuredfor selective operation in (i) the first mode to determine an aboveground range from the portable device to the transmitter based on anabove ground measured signal strength of the electromagnetic signal anda surface effect compensation, and (ii) the second mode with thetransmitter in the ground to determine a depth of the transmitter belowthe surface based on a below ground measured signal strength of theelectromagnetic signal.

In yet another aspect of the disclosure, a system and associated methodfor use in horizontal directional drilling are described, the systemincludes a drill string that extends from a drill rig to an ingroundtool such that extension and retraction of the drill string moves theinground tool through the ground during an inground operation. Thesystem includes a transmitter that includes one or more sensors formeasuring one or more operational parameters characterizing the statusof the inground tool, wherein the transmitter transmits at two or morefrequencies with at least one of such frequencies itself representingmultiple data bits characterizing the status of the inground toolirrespective of any modulation of the frequencies. A portable device, asanother part of the system, includes a receiver configured to receivethe two or more frequencies and a processor configured to recover thestatus of the inground tool from the two or more frequencies. Anassociated portable device, transmitter and methods are also described.

BRIEF DESCRIPTIONS OF THE DRAWINGS

Example embodiments are illustrated in referenced figures of thedrawings. It is intended that the embodiments and figures disclosedherein are to be illustrative rather than limiting.

FIG. 1 is a diagrammatic view of an embodiment of a system forperforming an inground operation in accordance with the presentdisclosure using an advanced communication protocol and methods.

FIG. 2 is a diagrammatic, partially cutaway view, in perspective, whichillustrates an embodiment of a transmitter produced in accordance withthe present disclosure.

FIG. 3 is a block diagram illustrating additional details of thetransmitter with respect to the embodiment of FIG. 2.

FIG. 4 is a block diagram illustrating details of an embodiment of afrequency synthesizer which forms part of the embodiment of thetransmitter of FIGS. 2 and 3.

FIG. 5a is a diagrammatic representation of an embodiment of frequencybands and sub-bands based on which various embodiments of a transmitteraccording to the present disclosure can be configured.

FIG. 5b is a diagrammatic representation of an embodiment of a set oflookup tables that can be used for the depth and symbol frequency lookuptables shown in FIG. 4.

FIGS. 5c-5j are diagrammatic representations of embodiments of thelook-up tables in the set of look-up tables of FIG. 5b , shown here toillustrate further details with respect to each look-up table.

FIG. 6a is a plot of the power spectral density of noise taken at a highresolution, corresponding to an actual physical location at which a 50Hz powerline frequency is in use.

FIG. 6b is a diagrammatic illustration of one embodiment of a screenshot showing a display 36 including a bar graph display illustrating theaverage noise per frequency sub-band for the embodiment of sub-bandsinitially shown in FIG. 5 a.

FIG. 7 is a further enlarged view of sub-band 10 from FIG. 6a , shownhere to facilitate a discussion of the section of a depth frequency andsymbol frequencies and including an inset view that illustrates anembodiment of the spectral content of symbols transmitted in accordancewith the present disclosure.

FIG. 8 is a flow diagram that illustrates an embodiment for theoperation of a transmitter according to the present disclosure.

FIG. 9 is a block diagram illustrating an embodiment of the portabledevice shown in FIG. 1.

FIG. 10a is an expanded view of frequency sub-band 6 from FIG. 6 a.

FIG. 10b is a diagrammatic illustration of one embodiment of a screenshot illustrating the appearance of sub-band 6 on a display screen forpurposes of operator selection and modification of symbol frequencies,as well as other functions in accordance with the present disclosure.

FIG. 11 is a further enlarged view of a portion of sub-band 6 of FIGS.10a and 10b , shown here for purposes of describing further details withrespect to symbol frequency selection.

FIG. 12 is a flow diagram illustrating an embodiment of a method foroperating a portable device in accordance with the present disclosurefor purposes of spectral scanning and symbol frequency assignment.

FIG. 13 is a flow diagram illustrating an embodiment of a method foroperating a portable device in accordance with the present disclosure ina normal mode during an inground operation.

FIG. 14 is a block diagram illustrating an embodiment of a depthdetector for determining the depth of the transmitter based on receptionof symbol frequencies in accordance with the present disclosure.

FIG. 15 is a block diagram illustrating another embodiment of a depthdetector for determining the depth of the transmitter based on receptionof symbol frequencies in accordance with the present disclosure.

FIG. 16 is a block diagram illustrating yet another embodiment of adepth detector for determining the depth of the transmitter by using anerror correction code to correct amplitudes associated with receivedsymbols.

FIG. 17 is a flow diagram illustrating an embodiment of a method fordepth determination based on the reception of symbol frequencies, inaccordance with the present disclosure.

FIG. 18 is a diagrammatic illustration of an embodiment of an antennafor use in the portable device of FIG. 1.

FIG. 19 is a diagrammatic illustration of a symbol stream including asilent sync implementation, in accordance with the present disclosure.

FIG. 20 is a diagrammatic plot of a transmitted data waveform, shownhere to illustrate further details with respect to silent sync, inaccordance with the present disclosure.

FIG. 21 is a diagrammatic further enlarged plot of a portion of areceived signal based on the transmitted data stream of FIG. 20, shownhere subject to noise.

FIG. 22 is a diagrammatic plot illustrating a waveform that is thesquare of the waveform of FIG. 21 such that a null symbol is readilyidentifiable.

FIG. 23 is a flow diagram illustrating an embodiment of a method foroperating the system of the present application using silent sync.

FIG. 24 is a plot that diagrammatically illustrates a surface effecterror versus distance in accordance with the present disclosure.

FIG. 25 is a diagrammatic illustration of an embodiment of theappearance of the display screen of a portable device operating in anabove ground range test mode in accordance with the present disclosure.

FIG. 26 is a diagrammatic illustration of an embodiment of theappearance of the display screen of a portable device operating in anormal mode in accordance with the present disclosure.

FIG. 27 is a flow diagram illustrating an embodiment of a method foroperating a portable device in a dual mode configuration including anabove ground range test mode and a normal mode in accordance with thepresent disclosure.

FIG. 28 is block diagram illustrating an embodiment of a receiversection, forming part of a portable device, which receives a multi-bitsymbol stream in time slices for transfer to a spectrogram buffer andwhich excludes time slices from the spectrogram buffer that aredetermined as exceeding a noise threshold in accordance with the presentdisclosure.

FIG. 29 is a flow diagram illustrating an embodiment of a method forloading spectrogram time slices into spectrogram buffer and forsubsequently decoding the time slices to recover packet data inaccordance with the present disclosure.

FIG. 30 is a diagrammatic illustration showing the contents of thespectrogram buffer and details relating to decoding the contents inaccordance with the present disclosure.

FIG. 31 is a flow diagram illustrating an embodiment of a decode processfor recovering packet data from the spectrogram buffer of FIG. 30 inaccordance with the present disclosure.

DETAILED DESCRIPTION

The following description is presented to enable one of ordinary skillin the art to make and use the invention and is provided in the contextof a patent application and its requirements. Various modifications tothe described embodiments will be readily apparent to those skilled inthe art and the generic principles taught herein may be applied to otherembodiments. Thus, the present invention is not intended to be limitedto the embodiment shown, but is to be accorded the widest scopeconsistent with the principles and features described herein includingmodifications and equivalents. It is noted that the drawings are not toscale and are diagrammatic in nature in a way that is thought to bestillustrate features of interest. Descriptive terminology may be adoptedfor purposes of enhancing the reader's understanding, with respect tothe various views provided in the figures, and is in no way intended asbeing limiting.

A bit, for purposes of the present application, is a binary data valuehaving two states characterized such as 1/0, +/−, and the like. Asymbol, for purposes of the present disclosure, is a data value thatrepresents one or more bits. A multi-bit symbol represents two or morebits. A symbol can characterize any suitable type of information suchas, for example, pitch data, roll data, temperature data, battery dataand synchronization data, without limitation. Different multi-bitsymbols represent different, multi-bit data values. For example, 16different symbols can represent a four bit data value. Each multi-bitsymbol, for purposes of the present disclosure, is represented by adistinct frequency that is different from the frequency that isassociated with any other multi-bit symbol. A symbol stream is made upof a serial transmission of multi-bit symbols such that the symbolstream is decodable into a corresponding digital data stream, which canbe binary. The symbol stream can be transmitted subject to a packetstructure such that the particular position of a given symbol within thepacket structure defines a data type that is associated with thatsymbol.

Turning now to the drawings, wherein like items may be indicated by likereference numbers throughout the various figures, attention isimmediately directed to FIG. 1, which illustrates one embodiment of asystem for performing an inground operation, generally indicated by thereference number 10. The system includes a portable device 20 that isshown being held by an operator above a surface 22 of the ground as wellas in a further enlarged inset view. It is noted that only limitedinter-component cabling is shown within device 20 in order to maintainillustrative clarity, but all necessary cabling is understood to bepresent and may readily be implemented by one having ordinary skill inthe art in view of this overall disclosure. Device 20 includes athree-axis antenna cluster 26 measuring three orthogonally arrangedcomponents of magnetic flux. One embodiment of a useful antenna clustercontemplated for use herein is disclosed by U.S. Pat. No. 6,005,532which is commonly owned with the present application and is incorporatedherein by reference. Details with respect to the embodiment of theantenna utilized herein will be provided at an appropriate pointhereinafter. Antenna cluster 26 is electrically connected to anelectronics section 32. A tilt sensor arrangement 34 may be provided formeasuring gravitational angles from which the components of flux in alevel coordinate system may be determined. An appropriate tilt sensorincludes, by way of non-limiting example, a triaxial accelerometer.

Device 20 can further include a graphics display 36 and a telemetryantenna 40. The latter can transmit or receive a telemetry signal 44 fordata communication with the drill rig. It should be appreciated thatgraphics display 36 can be a touch screen in order to facilitateoperator selection of various buttons that are defined on the screenand/or scrolling can be facilitated between various buttons that aredefined on the screen to provide for operator selection. Such a touchscreen can be used alone or in combination with an input device 48 suchas, for example, a trigger button. The latter can be used without theneed for a touch screen. Moreover, many variations of the input devicemay be employed and can use scroll wheels and other suitable forms ofselection device either currently available or yet to be developed. Theelectronics section can include components such as, for example, one ormore processors, memory of any appropriate type, antenna drivers andanalog to digital converters. As is well known in the art, the lattershould be capable of detecting a frequency that is at least twice thefrequency of the highest frequency of interest. Other components may beadded as desired such as, for example, a magnetometer 50 to aid inposition determination relative to the drill direction and ultrasonictransducers for measuring the height of the device above the surface ofthe ground.

Still referring to FIG. 1, system 10 further includes drill rig 80having a carriage 82 received for movement along the length of anopposing pair of rails 84. An inground tool 90 is attached at anopposing end of a drill string 92. By way of non-limiting example, aboring tool is shown as the inground tool and is used as a framework forthe present descriptions, however, it is to be understood that anysuitable inground device may be used such as, for example, a reamingtool for use during a pullback operation or a mapping tool. Generally,drill string 92 is made up of a plurality of removably attachable drillpipe sections such that the drill rig can force the drill string intothe ground using movement in the direction of an arrow 94 and retractthe drill string responsive to an opposite movement. The drill pipesections can define a through passage for purposes of carrying adrilling mud or fluid that is emitted from the boring tool underpressure to assist in cutting through the ground as well as cooling thedrill head. Generally, the drilling mud also serves to suspend and carryout cuttings to the surface along the exterior length of the drillstring. Steering can be accomplished in a well-known manner by orientingan asymmetric face 96 of the boring tool for deflection in a desireddirection in the ground responsive to forward, push movement which canbe referred to as a “push mode.” Rotation or spinning 98 of the drillstring by the drill rig will generally result in forward or straightadvance of the boring tool which can be referred to as a “spin” or“advance” mode.

The drilling operation can be controlled by an operator (not shown) at acontrol console 100 which itself includes a telemetry transceiver 102connected with a telemetry antenna 104, a display screen 106, an inputdevice such as a keyboard 110, a processing arrangement 112 which caninclude suitable interfaces and memory as well as one or moreprocessors. A plurality of control levers 114, for example, controlmovement of carriage 82. Telemetry transceiver 104 can transmit orreceive a telemetry signal 116 to facilitate bidirectional communicationwith portable device 20. In an embodiment, screen 106 can be a touchscreen such that keyboard 110 may be optional.

In an embodiment, device 20 is configured for receiving anelectromagnetic depth signal 120 and an electromagnetic data signal 122that are transmitted from a transmitter 130 that is supported within theboring tool or other inground tool. These signals may be referred tocollectively herein as the transmitter signals. The transmitter signalscan be dipole signals. It should be appreciated that the portable devicecan be operated in either a walkover locating mode, as illustrated byFIG. 1, or in a homing mode having the portable device placed on theground, for example, as illustrated by U.S. Published Patent Applicationno. 2013/0175092 which is incorporated by reference in its entirety.While the present disclosure illustrates a dipole locating fieldtransmitted from the boring tool and rotated about the axis of symmetryof the field, the present disclosure is not intended as being limitingin that regard.

Information carried by the data signal can include, but is not limitedto position orientation parameters based on pitch and roll orientationsensor readings, temperature values, pressure values, battery status,tension readings in the context of a pullback operation, and the like.Device 20 receives the transmitter signals using antenna array 26 andprocesses received data signal 122 to recover the data, as will befurther described.

FIG. 2 is a diagrammatic, partially cutaway view, in perspective, whichillustrates an embodiment of transmitter 130. The latter includes a mainhousing 134 that can be at least generally cylindrical in configuration.A battery compartment 138 can be formed at one end of the housing withan opposing end 140 supporting a main printed circuit board (PCB) 144which itself can support an antenna 148 that emits the transmittersignals. An accelerometer module 150 can be positioned adjacent to oneend of PCB 144. Other sensors and components can be located on the mainprinted circuit board, as will be further described.

Attention is now directed to the block diagram of FIG. 3 in conjunctionwith FIG. 2 for purposes of describing additional details with respectto an embodiment of transmitter 130. The transmitter includes aprocessing section 152 that receives sensor information via amultiplexer 154. The multiplexer can be interfaced with any number ofsensors forming a sensor suite. In the present example, the sensorsinclude accelerometers 158 that are supported in accelerometer module150 of FIG. 2, a pressure sensor 160 which can be used to sense theannular pressure within the borehole around the transmitter, atemperature sensor 164, a battery current sensor 168 and a batteryvoltage sensor 170. External communication for the transmitter can beprovided, in some embodiments, by an external communication connection174. Such communication is not required to be transmitted through theground but rather can be performed while the transmitter is aboveground, for example, in a position adjacent to device 20. The externalcommunication can be implemented in any suitable manner including butnot limited to IrDA, NFC, Wi-Fi, Zigbee or Bluetooth. A power supplysection 178 can comprise a battery 180 that provides power via anovervoltage and reverse polarity detector 184. The latter provideselectrical power to a logic and sensor power supply 188 and to anantenna drive power supply 190. The logic and sensor power supplyprovides power to the sensor suite as well as to processing section 152.The antenna drive power supply feeds electrical power to a depth antennadriver 194 and a data antenna driver 198 which electrically driveopposing ends of an antenna coil forming part of antenna 148. Drivers194 and 198, in an embodiment, can be half bridge drivers. The antennadrivers receive input signals from a processor 200 that forms part ofthe processing section. The processing section further includes anoscillator 210 such as, for example, a crystal oscillator. Theoscillator can be selected to provide a relatively high degree oftemperature and overall stability. Processor (CPU) 200 includes a timersection 212 that can serve to generate a reference signal having astability that reflects the stability of oscillator 210. The outputfrequency of the timer is selectable based on a reload timer value thatcan be specified by the user. The processor is in data communicationwith a memory 218 which can include any suitable information including,but not limited to depth frequency information 224 and symbol frequencyinformation 228, each of which will be described at an appropriatepoints hereinafter.

Turning to FIG. 4, an embodiment of a frequency synthesizer is generallyindicated by the reference number 300 and is implemented as part ofprocessing section 152 of FIG. 3. It should be appreciated that thefrequency synthesizer can be implemented in hardware, software or anysuitable combination thereof. The embodiment of FIG. 4 is a two channeldirect digital synthesizer (DDS) having a depth channel 304 and a symbolchannel 308. The depth channel provides an output signal 310 to depthdriver 194 of FIG. 3 for producing depth signal 120 while the symbolchannel provides an output signal 312 to data driver 198 of FIG. 3 forproducing data signal 122 (FIGS. 1 and 2). A depth channel waveformlookup table section 320 and a symbol channel waveform lookup tablesection 324 each includes at least one waveform or phase lookup tablethat characterizes one period of a selected waveform such as, forexample, a sinusoid. In another embodiment, each of the depth channellookup table section and the symbol channel lookup table section caninclude a plurality of waveform or phase lookup tables. In the presentexample, there are n waveform lookup tables diagrammatically shown andindicated by the reference numbers 326 a-n. It should be appreciatedthat any desired waveform or waveforms can be characterized by thelookup tables. Further, there is no requirement for the depth channellookup table(s) and the symbol channel lookup table(s) to characterizethe same waveform(s). Each waveform lookup table 326 a-n can include alarge number of samples of the magnitude of the characterized waveformbased, for example, on the amount of memory that is available and thedesired resolution. The samples are selectively addressable by a depthchannel phase accumulator 330 and a symbol channel phase accumulator334, respectively, using an m-wide addressing arrangement. Each phaseaccumulator is configured to provide an output count to its respectivewaveform lookup table section based on an input increment or offset sizethat is provided by a depth channel frequency control 338 and a symbolchannel frequency control 340, respectively. In the present embodiment,the particular one of the waveform lookup tables 326 a-n to be used atany given time for each of the depth channel and the symbol channel isbased on the frequency to be generated, as will be further described.Each phase accumulator generates what can be described as a quantizedsawtooth waveform output that changes from one level or count to thenext by a respective one of the input increment sizes. In response toeach respective phase accumulator input count, the depth channel lookuptable that is currently in use and the symbol channel lookup table thatis currently in use sequentially generate digital output magnitudes thatare received by a depth channel pulse width modulator (PWM) generator350 and a symbol channel pulse width modulator (PWM) generator 352,respectively, on an n-wide address arrangement. Based on the magnitudevalue received by each PWM generator, a pulse width modulator generatesan output pulse train having an at least generally constant outputmagnitude but with a pulse width that increases in proportion to theoutput magnitude value from each lookup table. Filtering, via theinductive properties of antenna 148, smooths the waveform to approximatea desired output waveform such as, for example, a sinusoidal waveform.

Still referring to FIG. 4, each of a depth channel output waveform 360and a symbol channel output waveform 362 can be generated, for example,across a frequency range approaching 0 Hz to 45 KHz with a high degreeof accuracy. It should be appreciated that any suitable frequency rangecan be utilized and the range of 0 to 45 KHz has been described by wayof example and is not intended to be limiting. In the presentembodiment, the accuracy can be at least approximately +/−0.1 Hz or lessat a resolution of at least approximately 5 Hz. It is noted that thespecified accuracy, in the context of the present embodiment, is givenfor at least approximately 45 KHz which represents a lower limit onaccuracy across the frequency range. As compared to prior artapproaches, it should be appreciated that the present disclosureprovides for higher precision, greater consistency and remarkableflexibility with respect to frequency placement across the entiretransmission bandwidth. Output frequencies 360 and 362 are establishedbased on the input increment size provided to depth channel phaseaccumulator 330 via depth channel frequency control 338 and symbolchannel phase accumulator 334 via symbol channel frequency control 340.Depth channel frequency control 338 receives a depth frequency input 368that specifies the depth frequency. The depth channel frequency controlcan convert a specified depth frequency to an increment size for depthchannel phase accumulator 330 in any suitable manner. In an embodiment,the depth channel frequency control can include an increment lookuptable 370 that indexes depth frequency against the increment size. Inanother embodiment, a formula can be used to determine the incrementsize, as follows:

$\begin{matrix}{{{increment}\mspace{14mu} {size}} = \frac{\left( {{desired}\mspace{14mu} {frequency}} \right) \times \left( {{phase}\mspace{14mu} {accumulator}\mspace{14mu} {size}} \right)}{\left( {{phase}\mspace{14mu} {accumulator}\mspace{14mu} {update}\mspace{14mu} {rate}} \right)}} & \left( {{EQN}\mspace{14mu} 1} \right)\end{matrix}$

Where the phase accumulator size is chosen to provide the minimumrequired frequency resolution and the phase accumulator update rate isestablished by timer 212 (FIG. 3). Similarly, the symbol channelfrequency control can convert a specified symbol frequency received on adata symbol stream input 374 to an increment size for symbol channelphase accumulator 334 in any suitable manner such as, for example, byusing an increment lookup table 372 or a formula. The origin of the datasymbol stream for data symbol stream input 374 will be described at anappropriate point hereinafter. It is noted that there is no requirementfor the depth and symbol channel frequency controllers to use anidentical increment size lookup table. Table 1 below illustrates aportion of increment lookup table 372.

TABLE 1 Desired Output Frequency vs. Phase Accumulator Size IncrementDesired Output Frequency Phase Accumulator Increment (Hz) (counts) 5 150 10 500 100 32770 6554 45000 9000

Based on Table 1, it should be appreciated that a high degree ofresolution is provided in terms of the frequency that is selectable foreach of depth output frequency 360 and symbol output frequency 362. Inthe present embodiment, a resolution of 5 Hz can be provided across theentire frequency range extending from worldwide AC powerline frequenciesto 45 KHz. Of course, other embodiments can utilize a like or differentresolution to even higher frequencies. Other resolutions can be used,some of which are larger and some of which are even more fine, however,Applicants recognize that 5 Hz represents a relatively small commonmultiple of 50 Hz and 60 Hz which are the predominant powerlinefrequencies around the world. Further discussions with respect topowerline frequencies will be presented below.

With continuing reference to FIG. 4, it should be appreciated that depthoutput frequency 360 and symbol output frequency 362 are illustrated asfrequency tones that are of a limited or fixed duration, an at leastessentially fixed frequency and can include a variable magnitude.Magnitude/amplitude shaping can be accomplished using a depth channelwaveform/amplitude control 380 for the depth channel which may bereferred to as a depth channel shaper and a symbol channelwaveform/amplitude control 382 which may be referred to as a symbolchannel shaper. Another example output of depth channel PWM generator350 is a continuous depth signal 386 which is of at least essentially acontinuous magnitude. In this instance, depth channel shaper 380 may notbe needed, although it should be understood that its operation reflectsthe operation of the symbol channel shaper, as described herein. Itshould be appreciated that the depth of the transmitter, based on depthsignal 386, can be determined based on the well-known dipole equations,as described for example, in U.S. Pat. No. 5,633,589 which isincorporated herein by reference. Another example output 390 of symbolchannel PWM generator 352 illustrates a series of output symbolsindicated as 392 a-392 f which can vary in frequency from one symbol tothe next. As will be further described, output 390 can comprise a symbolstream. In the present embodiment, there is no gap or zero magnitudespace present or inserted between adjacent symbols by phase accumulator334. Thus, the frequency can change abruptly from one symbol to the nextin a way that can introduce noise responsive to such abrupt frequencytransitions. It should be appreciated that symbols 392 a-392 f areshaped in a way that avoids abrupt frequency transitions by beginningand ending at a value of approximately zero magnitude. Such shaping canbe accomplished through the application of a suitable window or taperingfunction by symbol channel shaper 382 such as, for example, a Hammingwindow, Hann window, Welch window or a triangular window, among others.What is common to all of the subject window functions resides in a zeromagnitude of the waveform for any point that is outside of a windowinterval such that each symbol starts and ends with a zero magnitudewaveform.

Attention is now directed to FIG. 5a in conjunction with FIG. 2.Although not a requirement, embodiments of transmitter 130 can beconfigured to transmit depth signal 120 and data signal 122 using aseries of transmitter bands, generally indicated by the reference number400 that extend from approximately 0 to 45 KHz. It should be understoodthat other embodiments can use different transmitter bands and sub-bandswith the present embodiment serving by way of a non-limiting example.While the value of zero is listed as a lower limit, it should beunderstood that the actual lower limit can be represented by worldwidepredominant power line frequencies or some higher value. The transmitterbands are indicated as BT1-BT5 and are also set forth in Table 2. Whilethe descriptive framework employed by Table 2 uses transmitter bandsthat include frequency sub-bands, it will become evident below that theconcept of transmitter bands is not generally applicable to embodimentsof a wideband transmitter, yet to be described, even though the termsub-band is considered to be applicable to a wideband transmitter in thesense of defining some limited portion of the overall bandwidth acrosswhich the wideband transmitter is capable of transmitting.

TABLE 2 Transmitter Bands and Sub-Bands Transmitter Band Sub-BandSub-Band Band Frequency Range No. Frequency Range BT1 0-4.5 KHz SB1    0 to 4.5 KHz BT2 4.5 KHz-9 KHz   SB2  4.5 KHz to 9 KHz BT3  9 KHz-18KHz SB3   9 KHz to 13.5 KHz SB4 13.5 KHz to 18 KHz BT4   18 KHz-31.5 KHzSB5   18 KHz to 22.5 KHz SB6 22.5 KHz to 27 KHz SB7   27 KHz to 31.5 KHzBT5 31.5 KHz-45 KHz   SB8 31.5 KHz to 36 KHz SB9   36 KHz to 40.5 KHz SB10 40.5 KHz to 45 KHz

Still referring to FIG. 5a , the frequency range from 0 to 45 KHz, inaccordance with the present embodiment, is further divided into 10sub-bands SB1-SB10, each of which is 4.5 KHz in width. Each band aboveBT1 and sub-band 1 can be considered as including its lower frequencylimit. It should be appreciated that any individual transmitter can beconfigured for transmission in one of transmitter bands BT1-BT5. The useof the transmitter bands, although not required, allows for matchingantenna 148 (FIG. 2) to the transmitter band such that transmissionefficiency is at least near optimal. While transmitter bands BT1 and BT2each include a single sub-band, it is noted that transmitter band BT3includes two sub-bands, SB3 and SB4, and transmitter bands BT4 and BT5each include three sub-bands: SB5-SB7 and SB8-SB10, respectively. Anembodiment of a transmitter according to the present disclosure can beconfigured to transmit depth signal 120 and data signal 122 in a singlesub-band. In another embodiment, a transmitter can be configured totransmit depth signal 120 in a sub-band that is different from thesub-band that is used for data signal 122. In this regard, it should beappreciated that the use of a separate synthesizer channel (FIG. 4) forthe depth channel provides for a great degree of flexibility with regardto the frequency of the depth signal in relation to the data signal. Instill another embodiment, a transmitter can be configured to transmit onmultiple sub-bands. For example, a transmitter configured to transmit ontransmitter band BT3 can transmit on both SB3 and SB4. As anotherexample, a wideband transmitter, as further described below, cantransmit on two or more sub-bands, such as SB4 and SB10 such that thesub-bands can even be spaced apart by other sub-bands. With regard tothe aforementioned wideband transmitter, which is described immediatelyhereinafter, it will be apparent that the concept of transmitter bandscan be thought of as inapplicable since a single antenna can be used forthe entire bandwidth of the transmitter.

In some embodiments, transmitter 300 can be configured to cooperate withantenna 148 such that transmitter 130 transmits over a wide frequencyrange or band extending from a lowermost frequency to approximately 45KHz or higher. In this way, this wide frequency band can be covered by asingle wideband transmitter, using a single antenna, while maintainingsuitable efficiency with respect to power consumption across the entirewide frequency range. It should be appreciated that, in the absence ofthe provisions described immediately hereinafter for at least part ofthe wide frequency range, an unmatched condition between the antenna andthe input frequency can produce unacceptable battery power consumptionto achieve the same RF output power.

Referring again to FIG. 4, in order to transmit across an entirefrequency range from a lowermost frequency to approximately 45 kHz, byway of non-limiting example, embodiments of depth channel lookup tablesection 320 and symbol channel lookup table section 324 can beconfigured to include sets of lookup tables 326 a-n. Any suitablenumber, n, of lookup tables can be used in each set. Depth channel phaseaccumulator 330 and symbol channel phase accumulator 334 can beconfigured to utilize the appropriate depth channel lookup table andsymbol channel lookup table, respectively, based on the frequency to begenerated. As will be described in further detail at an appropriatepoint hereinafter, each individual lookup table of these sets of lookuptables can be customized to drive antenna 148 in a way that maintainspower at an at least generally constant power consumption over a portionof the overall wide transmission bandwidth such that the combination oflookup tables maintains a desired level of power consumption over theentire wide transmission bandwidth. Each lookup table can be configuredfor driving the antenna not only based on providing a particular waveshape but also using a selected drive waveform magnitude. Accordingly,transmission power and transmitter power consumption can be controlledor regulated, at least in part, based on the magnitude of the lookuptable waveform. While some of the lookup tables in a set can be providedfor purposes of limiting and/or controlling power consumption, one ormore lookup tables can be provided for purposes of implementing a highoutput power mode. In such a case, a high output power lookup table canexhibit the same sampled waveform shape as a corresponding lower powerlookup table, but the high power lookup table includes an increasedmagnitude version of the sampled waveform. The antenna, in theembodiments presented herein, is not required to be driven at a resonantfrequency. In this regard, the resonant frequency that is presented bythe inductance of antenna 148, in combination with any parasiticcapacitances, is generally far higher than a highest frequency of thetransmission range such as, for example, 45 kHz. For instance, theresonant frequency can be in the megahertz range. In this regard, theantenna can exhibit an impedance across the transmission frequency rangethat can be considered as a constant, at least from a practicalstandpoint. Thus, antenna 148 can include a number of windings that isselected based, at least in part, on a selected or targeted amount ofcurrent draw from battery 180.

FIG. 5b illustrates an embodiment of a set of lookup tables, generallyindicated by the reference number 450, that can be used for the depthand symbol frequency lookup tables 326 a-n. In this case, the setcomprises 8 lookup tables. It should be noted that there is norequirement to use the same set of lookup tables for depth and datatransmission. In this embodiment, the lookup table set covers SB3through SB10, corresponding to a wideband frequency range of 9 KHz to 45KHz. While the wide frequency range is characterized in terms ofsub-bands for purposes of descriptive continuity, it should beappreciated that there is no need for frequency confinement based on thepreviously described transmitter bands and/or sub-bands (see, forexample, Table 2) in the context of a wideband transmitter. FIG. 5billustrates general lookup table waveform shapes 460, 462, 464, 466,468, 470, 472 and 474 that are used for each sub band SB-3-SB10,respectively. For SB3-SB6, a sinusoidal sampled waveform is used. ForSB7-SB10, a stepped sampled waveform is utilized. Further details willbe provided immediately hereinafter.

FIGS. 5c-5j , illustrate further enlarged plots of lookup table sampledwaveforms 460, 462, 464, 466, 468, 470, 472 and 474, respectively, forthe present embodiment of the set of lookup tables for purposes of depthand data transmission. The horizontal axis of each of these figuresillustrates the sampled waveform period or time slot, while the verticalaxis designates a Pulse Width Modulation percentage. It is noted thatthe actual duration of the sampled waveform period is limited to 0-15 onthe time slot axis shown in each of these figures. In FIG. 5c , waveform460 includes a sinusoidal shape having a PWM percentage that ranges fromapproximately 15% PWM to 85% PWM. In FIG. 5d , waveform 462 includes asinusoidal shape having a PWM percentage that ranges from approximately10% PWM to 90% PWM. In FIG. 5e , waveorm 464 includes a sinusoidal shapehaving a PWM percentage that ranges from approximately 5% PWM to 95%PWM. In FIG. 5f , waveform 466 includes a sinusoidal shape having a PWMpercentage that ranges from approximately 0% PWM to 100% PWM. Thus, themagnitude of the sampled waveform increases by approximately 10%progressively through the subject figures until 100% PWM modulation isreached at waveform 466. Starting, however, with sampled waveform 468 ofSB7 in FIG. 5g , the sampled waveform changes dramatically. Inparticular, a stepped sampled waveform is utilized wherein the waveformtransits from 100% PWM to 0% PWM. For waveform 468, the on-time of thewaveform is approximately 20%. Thus, the use of sampled waveform 468generates a pulse train as an antenna drive signal with an on-time ofapproximately 20%. Referring to FIG. 5h , for waveform 470 whichcorresponds to SB8, the on-time of the waveform is approximately 27%.Thus, the use of sampled waveform 470 generates a pulse train as anantenna drive signal with an on-time of approximately 27%. Referring toFIG. 5i , for waveform 472 which corresponds to SB9, the on-time of thewaveform is approximately 33%. Thus, the use of sampled waveform 470generates a pulse train as an antenna drive signal with an on-time ofapproximately 33%. Referring to FIG. 5j , for waveform 474 whichcorresponds to SB10, the on-time of the waveform is approximately 50%corresponding to a square wave. Thus, the use of sampled waveform 470generates a pulse train as an antenna drive signal with an on-time ofapproximately 50%.

Referring collectively to the set of lookup tables of FIGS. 5c-5j , itshould be appreciated that this embodiment has been developed forpurposes of power control such that the transmitter draws or consumesapproximately the same amount of power irrespective of the specifictransmission frequency within a wide bandwidth. In this regard, given aconstant drive voltage and waveform, the transmitter would otherwisedraw increasingly more power as the frequency is reduced. Accordingly,the descriptions which follow will consider the lookup table setbeginning from the upper end of the transmitter bandwidth.

Lookup table waveform 474 of FIG. 5j for SB10, from 40.5 KHz to 45 KHz,drives the antenna using a square wave at the fundamental of thefrequency that is to be transmitted. In doing so for the higherfrequencies, the amplitude of the fundamental, as a first harmoniccomponent of the square wave, is higher by about 2 dB than the amplitudeof a pure sinewave of a corresponding power. As the transmit frequencyreduces from SB9 to SB8, FIGS. 5i and 5h demonstrate that the on-time ofwaveforms 472 and 470 progressively decreases. Accordingly, as thetransmit frequency becomes lower, the drive waveform becomes morepulse-like to progressively lower the amount of energy at thefundamental frequency of the pulse train. By progressively narrowing thepulses in the pulse train, the power that is drawn by the transmitter iscompensated and does not significantly increase with decreasing transmitfrequency. This pulse train drive signal approach is employed until asinewave drive signal of at least approximately full magnitude matchesthe available transmit power from a pulse train drive signal. Such acondition is satisfied at SB6 which utilizes sinewave-shaped lookuptable waveform 466. As the transmit frequency is lowered still furtherin the SB5 and SB4 lookup tables, the magnitude of the sinewave drivewaveform is further reduced in order to compensate for the tendency oftransmit power to increase responsive to decreasing transmit frequency.The lookup table approach developed by Applicants, which is submitted tobe heretofore unknown, provides for varying the drive frequency over awideband transmit range using a single antenna and without the need forusing different antennas which would necessitate the use of multi-coilantennas, complex antenna coil switching and/or complex variable drivevoltage arrangements.

Based on the foregoing, the present disclosure can provide a widebandtransmitter having a single antenna that is driven across a widefrequency band in a way that can maintain constant or controlled powerconsumption, at least to an approximation, when the power consumptionwould otherwise exhibit large variations across that same frequency bandby using a single drive signal waveform. Variation in the powerconsumption across the wide frequency band can be limited to acceptablylow levels across the range of 9 KHz to 45 kHz. In this way, Applicantsare able to provide a wideband transmitter that operates across a widefrequency range with power consumption regulation and control that issubmitted to have been unseen heretofore. In the past, performinginground operations at different frequencies for depth and locating dataoften required the purchase of a transmitter that was dedicated to eachfrequency of interest. The recognitions that have been brought to lightherein can result in significant cost savings since a single widebandtransmitter can replace a plurality of prior art transmitters. In thisregard, the teachings herein are equally applicable with respect to atransmitter that transmits a depth frequency or tone at one discretefrequency and transmits a data signal at a different frequency that ismodulated in any suitable manner such as, for example, using BPSK, QPSKor Manchester encoding.

With reference to FIG. 4 and in an embodiment, the symbol channel can beset up to output a single carrier frequency, much like depth signal 386of the depth channel, and that carrier frequency can be modulated in anysuitable manner, for example, to carry sensor data based, at least inpart, on the set of lookup tables 326. It should be appreciated thatthere are benefits associated with transmitting the depth frequency ortone at a relatively low frequency such as, for example, 1.5 kHz andtransmitting a modulated data frequency at a much higher frequency suchas, for example, in the range of 30 kHz to 45 kHz. Thus, a factor of 20or more can be provided between the modulated carrier frequency and thedepth tone as a result of the remarkable frequency generationcapabilities of a transmitter that is produced in accordance with thepresent disclosure. In this regard, low depth tone frequencies areassociated with avoiding sources of passive interference such as rebarwhile higher data frequencies are associated with higher rates of datathroughput based on the Nyquist rate. The present disclosure allows forthe transmission of a depth tone that is spaced apart from a modulateddata frequency by an amount that is submitted to be heretofore unseen,particularly when a single antenna is used to transmit both. Forexample, the depth tone can be transmitted at 1.5 kHz or less and themodulated data frequency can be transmitted in the range from 30 kHz to45 kHz. In another embodiment, 10 kHz can be used for the depth tonewhile 40 kHz can be used for the modulated data frequency.

Having described in detail above transmitters and associated componentsaccording to the present disclosure, details with respect totransmission of data signal 122 will now be brought to light. Inparticular, an M(ary) frequency shift keying approach is used such thata plurality of different symbols can be streamed to make up data signal122. In an embodiment, the data signal can serve to transmit a multi-bitsymbol stream. The ability to transmit a multi-bit symbol stream isfacilitated, at least in part, based on the use of synthesizer 300 ofFIG. 4. In particular, a multi-bit data symbol stream can be provided atdata symbol stream input 374 to symbol channel frequency control 340. Inthis way, data symbols corresponding to a wide variety of distinctfrequencies can be specified as part of the data symbol stream with eachdifferent symbol corresponding to a different frequency. In anembodiment, the data symbols of the symbol stream can correspond to 16symbols (4 bits), although any suitable number of symbols can be used,based on a desired data throughput. FIG. 4 illustrates output 390 basedon 16 symbols, S0-S15, with S0 corresponding to a lowest frequency andeach successively higher-numbered symbol corresponding to a relativelyhigher frequency, although this is not required and the mapping orassignment of symbols to frequencies can be performed in any suitablemanner. Thus, output 390 corresponds to an example input symbol streamof S2, S12, S2, S15, S0 and S10 at input 374.

FIG. 6a is a plot of the power spectral density of noise taken at a highresolution, generally indicated by the reference number 500,corresponding to an actual physical location at which a 50 Hz powerlinefrequency is in use. The signal level is shown on the vertical axis andthe frequency is shown on the horizontal axis. The frequency range of 0to 45 KHz corresponds to the frequency range that is covered by therange of transmitters described in accordance with the presentdisclosure, for example, with regard to FIG. 5a . As noted above, thepresent embodiment, utilizing the range of 0 to 45 KHz, has beenprovided by way of non-limiting example. Transmitter sub-bands SB1-SB10are also indicated. An initial choice of which sub-band is most suitablecan be based on a determination of an average noise value per sub-band.On this basis, any one of sub-bands SB8-SB10 appears to represent anacceptable choice while one of sub-bands SB1-SB3 appears to representthe worst choice.

While the spectral scan of FIG. 6a illustrates spectral informationessentially at a single location, it should be appreciated that spectralinformation can be collected in a cumulative manner. For example,spectral scanning can be performed while an operator walks the plannedborepath with device 20 while the device characterizes the noiseenvironment. In this way, the spectral plot of FIG. 6a can be thought ofas representing the noise environment along the entire planned borepathwith subsequent frequency selections being based on the noiseenvironment as characterized for the entire length of the plannedborepath while still utilizing the frequency selection techniques thathave been brought to light herein.

FIG. 7 is a further enlarged view of sub-band 10 from FIG. 6a ,generally indicated by the reference number 550, and is shown here toillustrate the selection of a depth frequency and sixteen symbolfrequencies S0-S15 within this sub-band. Each selected frequency hasbeen designated by an arrow. The various frequencies have been selected,for example, based on their correspondence to low noise points in thenoise plot. Based on the selection of frequencies, such as S0-S15 eitherautomatically and/or manually, Applicants submit that system 10 canprovide a level of noise immunity that has heretofore been unseen withrespect to performing an inground operation such as, for example,horizontal directional drilling and related pull-back or back-reamingoperations. Related considerations and further details will be providedin the context of a discussion of device 20 which receives the depthsignal and the data signal and which also can assist in theidentification of the depth signal frequency and symbol frequencies tobe used by the transmitter.

Attention is now directed to considerations with regard to powerlineharmonics that are frequently encountered in an ambient noiseenvironment. It should be appreciated, however, that frequency selectionbased on attempting to avoid powerline harmonics is not a requirementherein. That is, frequency selection based on low noise measurements,potentially in conjunction with other statistical noise characterizationparameters can provide remarkable benefits with respect to providingnoise immunity. Nonetheless, Applicants recognize that detailedexamination of noise plots, such as the one illustrated in FIG. 6a , atleast generally reveal the presence of powerline harmonics that arespaced apart by an increment that is established by the local powerlinefrequency. In the instance of a 60 Hz powerline frequency, the harmonicsare at least generally spaced apart by 60 Hz and can extend to valuesupwards of 30 KHz. Similarly, in the instance of a 50 Hz powerlinefrequency, the harmonics are at least generally spaced apart by 50 Hzand can extend to values upwards of 30 KHz. Therefore, powerlineharmonic noise can be reduced by choosing symbol frequencies that fallat least between or halfway in between the powerline harmonics. Giventhe assumption of stability in the powerline frequency, for a 50 Hzpowerline frequency, symbol frequencies in the series 75 Hz, 125 Hz, 175Hz, 225 Hz, and so on can be selected while, for a 60 Hz powerlinefrequency, symbol frequencies in the series 90 Hz, 150 Hz, 210 Hz, 270Hz, and so on can be selected. The subject symbol potential frequenciesmay be referred to herein as the in-between frequencies. As discussedabove, synthesizer 300 can be configured with a frequency resolution of5 Hz such that any desired in-between harmonic frequency of 50 Hz or 60Hz can be selected as a symbol frequency. Applicants recognize thatinstability in the powerline frequency, however, will cause a shift inthe harmonic frequencies. This shift increases with increasingfrequency. For example, if the fundamental powerline frequency isshifted by 0.1 Hz from 50 Hz, the 100th harmonic will shift by 10 Hz.Frequency selection, in view of such harmonic shifts, will be addressedat an appropriate point hereinafter.

Referring again to FIG. 7 and in an embodiment, synthesizer 300 can beconfigured to allow for frequency selection at a resolution of 5 Hz. Inthis regard, it should be appreciated that this level of resolutionprovides for frequency selection that is half way between adjacentpowerline harmonics. For a 50 Hz powerline frequency, the bandwidthbetween adjacent harmonics is at least approximately 50 Hz. For a 60 Hzpowerline frequency, the bandwidth between adjacent harmonics is atleast approximately 60 Hz. Due to the stability with which the symbolsare generated in conjunction with symbol shaping as seen in FIG. 4, thesymbols transmitted to make up the symbol stream of data signal 122 canexhibit limited spectral spreading. Further, the spectral spreading thatis present can exhibit a particular relationship with adjacent powerlineharmonics based on transmission rates. For example, a data throughput of50 bits per second requires transmission of 12.5 symbols per second for4 bit symbols for a symbol duration of 0.08 sec. As another example, adata throughput of 60 bits per second requires transmission of 15symbols per second for 4 bit symbols for a symbol duration ofapproximately 0.067 sec. Based on a 50 bit per second transmission rate(12.5 symbols per second), an inset view 552 in FIG. 7 includes a plot553 that illustrates the spectral content of symbol S13 in Hertzrelative to its nearest powerline harmonics. The fundamental frequencyof S13 is 43,825 Hz which is half way between adjacent 50 Hz powerlineharmonic frequencies of 43,800 Hz and 43,850 Hz. A fundamental peak 554is present in the spectral plot of the symbol at 43,825 Hz such that allof the spectral energy of this peak falls between the adjacent powerlineharmonics. Side lobes 556 a and 556 b also fall entirely between theadjacent powerline harmonics. It is noted that several additional sidelobes are shown having energy that falls outside of the adjacentpowerline harmonics at 43,800 Hz and 43,850 Hz. Remarkably, it should beappreciated that the spectrum of the symbol exhibits nulls 558 a and 558b that fall directly on the adjacent powerline harmonic frequencies.Thus, the spectral content of each symbol frequency effectively placesno signal power on the adjacent powerline harmonics. The nulls arepositioned to fall on the adjacent powerline harmonics, as shown in FIG.7, based on the data transmission rate of the symbol stream. Asdiscussed above, symbol stream 390 is transmitted without gaps betweenthe symbols. It should be appreciated that additional side lobes willlikewise be separated by nulls that are located directly on powerlineharmonic frequencies such as, for example, 43,750 Hz and 43,900 Hz,which have not been shown due to illustrative constraints. In thisregard, the symbol spectra includes a null positioned at every powerlineharmonic frequency. It is noted that an additional portion of thespectral energy that is associated with each symbol can be moved betweenadjacent powerline harmonics. For example, if the symbol transmissionrate is reduced by one half, additional side lobes shown in inset view552 of FIG. 7 will be positioned between the adjacent powerlineharmonics at 43,800 Hz and 43,850 Hz along with the fundamental peak. Atthe same time, nulls continue to fall directly on all of the powerlineharmonics. For the spectral plot of FIG. 7, it is noted thatapproximately 94 percent of the total energy associated with the symbolis contained by the fundamental and side lobes 556 a and 556 b.

Table 3 provides at least approximate values for each of the selectedfrequencies shown in FIG. 7. The reader is reminded that frequenciesS0-S15 were selected on the basis of exhibiting low noise, as opposed toattempting to avoid power line harmonics. In this regard, Table 3 alsolists a nearest powerline harmonic based on a 50 Hz powerline frequency.In some instances, such as with respect to the frequency choices for S1,S4, S7 and S8, it appears that these frequencies correspond to a 50 Hzpowerline harmonic while, in other cases, the frequency choices only forthe depth signal, S5 and S13 fall on in-between harmonic frequencies. Inthis regard, it should be appreciated that such shifting of low noisepoints in the noise spectrum can result from powerline frequency drift,as discussed above.

TABLE 3 EXAMPLE SELECTED FREQUENCIES Nearest Between DesignationFrequency (Hz) Harmonic Frequency Depth Signal 40,675 40,675 S0 40,74040,725 S1 40,850 40,825 or 40,875 S2 41,085 41,075 S3 41,210 41,225 S441,500 41,475 or 41,525 S5 41,825 41,825 S6 42,235 42,225 S7 42,40042,375 or 42,425 S8 42,700 42,675 or 42,725 S9 42,845 42,825 S10 43,20543,225 S11 43,420 43,425 S12 43,665 43,675 S13 43,825 43,825 S14 44,36044,375 S15 44,635 44,625

Based on the foregoing, Applicants submit that system 10 can provide alevel of noise immunity that has heretofore been unseen with respect toperforming an inground operation such as, for example, horizontaldirectional drilling and related pull-back or back-reaming operations.Related considerations and further details will be provided in thecontext of a discussion of device 20 which receives the depth signal andthe data signal and which also can assist in the identification of thedepth signal frequency and symbol frequencies to be used by thetransmitter. It should be appreciated that the depth signal frequencyand symbol frequency ordering given by Table 3 are not required. Thatis, the depth signal frequency can be positioned between symbolfrequencies. Based on the use of a separate channel for purposes ofgenerating the depth signal (FIG. 4), the depth signal can be positionedin a different sub-band than the symbol frequencies. Further, the symbolfrequencies can be reordered or rearranged in any suitable manner. Withregard to constraining frequency selection to a single sub-band, itshould be understood that an embodiment of a wideband transmitter can beconfigured to operate in a manner that mimics the operation of atransmitter that is constrained to operate based on sub-bands. Forexample, the selected frequencies in a wideband transmitter can belimited or constrained to a single sub-band, even though the widebandtransmitter is capable of transmission over a wide range of sub-bands.

FIG. 8 is a flow diagram that illustrates an embodiment for theoperation of a transmitter, generally indicated by the reference number600, according to the present disclosure. It is noted that, for purposesof the present discussion, it will be assumed that the depth frequencyas well as the frequencies associated with symbols S0-S15 have alreadybeen selected. These frequency choices can be stored at any suitablelocation such as, for example, in depth frequency table 224 and symbolfrequency table 228 of FIG. 3. The method begins at 604 and proceeds to608 which looks up the depth frequency increment, for example, fromlookup table 370 (FIG. 4) as part of the operation of depth channelfrequency control 338. In an embodiment that uses a single depth channelwaveform lookup table such as table 326 a in FIG. 4, depth channel phaseaccumulator 330 can always address that single waveform lookup table. Onthe other hand, in an embodiment that uses a plurality of depth waveformlookup tables, step 608 also can identify the correct waveform lookuptable 326 a-n as part of depth channel waveform tables section 320 inFIG. 4 such that depth channel phase accumulator 330 addresses theappropriate depth channel lookup table waveform based on frequency. Atstep 610, depth channel phase accumulator 330 receives the value andbegins counting based on the depth frequency increment, thereby causingthe appropriate depth channel lookup table 326 a-n and depth channel PWMgenerator 350 to begin continuously generating depth channel frequency386 to emit depth signal 120 at this frequency. At 614, CPU 200 readssensor information via multiplexer 154 to collect sensor data that is tobe transmitted. At 618, the CPU assembles the sensor data into a symbolstream which can invoke a packet structure that is yet to be described.The symbol stream is provided as data stream symbol input 374 to symbolchannel frequency control 340 in FIG. 4. At 620, the symbol channelfrequency control can utilize its lookup table 372 to identify theappropriate frequency for a current symbol to be transmitted. In anembodiment that uses a single symbol channel waveform lookup table suchas table 326 a in FIG. 4, symbol channel phase accumulator 334 canalways address that single waveform lookup table. On the other hand, inan embodiment that uses a plurality of symbol waveform lookup tables,step 620 also can identify the correct waveform lookup table 326 a-n aspart of symbol channel waveform tables section 324 in FIG. 4 such thatsymbol channel phase accumulator 334 addresses the appropriate symbolchannel lookup table waveform. It should be appreciated that thetransmission of a given symbol stream can necessitate that step 620switches data waveform lookup tables 326 a-n on a symbol-by-symbol basisfrom one symbol to the next, based on frequency. At 624, the currentsymbol is transmitted. Step 624 checks for the availability of anothersymbol to transmit. If a symbol is available, operation returns to 620such that the process repeats for the next symbol. On the other hand, ifthe next symbol is not yet ready, operation can return to 610 whichcontinues transmission of the depth signal. Sensor data is then againread at 614 and the process continues therefrom. It should beappreciated that data signal 122 is most often transmitted on anessentially continuous basis simultaneously with depth signal 120.

Having described embodiments of transmitter 130 in detail above,attention is now directed to FIG. 9 in conjunction with FIG. 1 forpurposes of describing additional details with respect to device 20which may be referred to interchangeably as a locator or receiver.Device 20 includes a battery 700 that feeds a power supply 704 whichsupplies appropriate electrical power to all of the components of thedevice, indicated as V+. Electronics section 32 includes a processor 710that is interfaced with a memory 714. A telemetry section 720 iscontrolled by the processor and coupled to antenna 40 for bidirectionalcommunication via signal 44. In some embodiments, the telemetry link canbe unidirectional from device 20 to the drill rig, in which casetransceiver 102 need only include receiver functionality. An externalcommunication arrangement 722 provides for external communication with atransmitter using external communication connection 174 (FIG. 3) of thetransmitter. As discussed above, such communication is not required tobe transmitted through the ground but rather can be performed while thetransmitter is above ground, for example, in a position adjacent todevice 20. The external communication can be implemented in any suitablemanner including but not limited to IrDA, NFC, Wi-Fi, Zigbee orBluetooth. A wide-band front end 730 is configured for receiving depthsignal 120 and data signal 122 using X, Y and Z antennas which make upantenna cluster 26 for measuring three orthogonal components of thesubject signals as well as for performing noise measurements along theseaxes, as is yet to be described. Additional details with respect to anembodiment of the antenna cluster will be provided at an appropriatepoint hereinafter. Each of the X, Y and Z antennas is interfaced to alow noise amplifier (LNA) 734 a, 734 b and 734 c, respectively, each ofwhich can be identically configured. The amplified output of each LNA issupplied to a respective one of filters 738 a, 738 b and 738 c, each ofwhich can be configured identically and which may be referred tocollectively as filters 738. Each filter serves as a bandpass filter 740exhibiting a low frequency roll-off or corner and a high frequencyroll-off or corner. While filters 738 are illustrated as individualfunctional blocks, it should be appreciated that the filters can beimplemented in any suitable manner. By way of non-limiting example, eachfilter can be implemented as a series of RC high-pass and low-passfilters that are distributed throughout the signal chain. In anembodiment, two high-pass filters can each be set at a low cornerfrequency of about 4 KHz and four low pass filters can be set at a highcorner frequency of about 90 KHz. This embodiment yields a relativelyflat frequency response from 10 KHz to 50 KHz. The roll-off below 10 KHzis approximately 40 dB of attenuation per decade and the roll-off above50 KHz is approximately 80 dB of attenuation per decade. It should beappreciated that the low end response of filters 738 and the low cornerfrequency can be established in consideration of the fundamental and loworder power line harmonics, which can be very strong. Amplifiers750a-750c can follow each respective one of filters 738 a-738 c withsufficient gain for purposes of driving each of analog-to-digitalconverters A/D 754 a-754 c. Each A/D 754 provides an output to CPU 710.In an embodiment, device 20 can be configured to receive the symbolstream in a way that suppresses powerline harmonic frequencies sincethere is effectively no energy present in the symbol stream at thepowerline harmonics. For example, the received signal can be processedsuch that the receiver response matches the symbol spectra asillustrated by plot 553 of FIG. 7. In particular, the spectral responseof the receiver can be matched to the spectral characteristics of thetransmitter by integrating the received symbol stream over a time periodthat corresponds to the time duration or period of each symbol. In thisway, the receiver frequency response matches the response of thetransmitter with respect to exhibiting null reception points at thepowerline harmonic frequencies. Accordingly, energy at the harmonicfrequencies is suppressed or ignored by the receiver while sweeping upthe spectral energy that is associated with the symbol. The receiver canemploy any suitable demodulation process that provides periodic nullsincluding but not limited to a Discreet Fourier Transform (DFT).

Still referring to FIG. 9 and having described an embodiment of locator20 in detail above, it should be appreciated that the locator can beconfigured for performing noise measurements and analysis for purposesof selecting a transmitter for transmission of the depth signal and thedata signal as well as establishing the frequencies to be associatedwith each of these signals. Of course, band selection may not berequired when a wideband transmitter is used. Noise measurements can bedetermined based on each orthogonal axis of antenna 26 (X, Y and Zantennas, as shown in FIG. 9). These individual noise components can beused to establish a three dimensional noise value, for example, based ona vector sum of the three antenna components. The vector sum can beuseful since the noise reading at a given point will essentially beinvariant with changes in the orientation of the locator. On the otherhand, displaying the noise reading obtained from a single axis willgenerally exhibit variation at a given point as the orientation of thelocator is changed. By allowing for monitoring noise along a singleantenna axis such as for example the X axis, an operator can identifywhich particular axis along the bore path may be problematic in terms ofinterference. Noise values can be determined in any suitable manner suchas, for example, based on a Fast Fourier Transform (FFT). In anembodiment, a noise scan can be produced from each axis for comparativepurposes. For example, an axis that exhibits relatively higher noisethan the other axes can be handled differently for purposes of datarecovery.

As discussed above with regard to FIG. 6a , an initial choice of whichsub-band is most suitable can be based, for example, on an average noisevalue per sub-band. In an embodiment, the locator can automatically makea recommendation to use the lowest average noise sub-band such as, forexample, SB-9. For example, display screen 36 can illustrate a plot, bargraph or any suitable form of display format based on the spectral scanof FIG. 6a , highlighting the selected sub-band. In some embodiments,the sub-band selection process can involve other statistical values forpurposes of characterizing the noise, as will be described immediatelyhereinafter.

FIG. 6b illustrates one embodiment of a screen shot showing display 36including a bar graph display illustrating the average noise persub-band wherein sub-band SB-10 is highlighted, for example, usinghatching and/or color or in some other suitable manner to indicate thatSB-10 has been automatically selected. In another embodiment, thelocator can make an automatic recommendation based on average noise persub-band in conjunction with other statistical values. Any suitablestatistical value(s) can be utilized including, for example, standarddeviation, minimum noise and peak noise. In still another embodiment,more than one sub-band can be recommended, in which case the user canselect between the recommended sub-bands. Recommending multiplesub-bands can be based on a limited amount of statistical variationbetween the sub-bands. For example, sub-bands 9 and 10 can both berecommended based on the relatively limited difference between the twosub-bands, as seen in FIG. 6b . As another example, multiple sub-bandscan be recommended, for instance, based on the average noise for a firstsub-band being lower than the average noise for a second sub-band whilethe peak noise for the first sub-band is higher than the peak noise forthe second sub-band. In an embodiment wherein more than one sub-band isrecommended, the system can be configured such that the user can selectone of such multiple recommended sub-bands for transmission. In anotherembodiment, the user can select multiple recommended sub-bands fortransmission. In yet another embodiment, one or more of such multiplerecommended sub-bands can be automatically selected for transmission.Since the information presented in FIG. 6a is based on a high resolutionnoise scan using a 5 Hz increment, a significant amount of noiseinformation can be extracted from the data. For example, the standarddeviation of the noise values within each sub-band can be determined.The heights of the various bars in FIG. 6b can be weighted by adding orsubtracting a value based on one or more other statistics. For example,if the standard deviation for a given sub-band is high, meaning that thenoise values are relatively more widely spread out, the height of theassociated bar can be maintained or even increased by some amount. Onthe other hand, if the standard deviation for a given sub-band is low,meaning that the noise values within that sub-band are relativelyconsistent, the height of the associated bar in FIG. 6b can be lowered.Similarly, the heights of the bars in the bar graph can be weightedbased on peak noise such that a sub-band having a high peak noise can beincreased in height by some amount. In any case, weighting can beperformed based on thresholds for the respective statistical values.Weighting can be applied based on individual statistical values orcombinations of statistical values. The automatically selected sub-bandcan be accepted by the operator touching an Auto-Select button 780 or bytouching any sub-band which he or she wishes to choose. The operator canoverride the automatic selection, for example, based on which specifictransmitters are currently available for performing the ingroundoperation. As another basis to present information to the operator,other statistical values can be presented. For example, overbars 781 (anumber of which are individually designated) show the peak noise persub-band. The operator may choose to avoid a sub-band that exhibits aparticularly high peak noise level, even if the average noise for thatsub-band is relatively low. For purposes of over-riding the automaticselection, the operator can touch a Manual Select button 782 and thentouch a sub-band which he or she wishes to choose. In anotherembodiment, display 36 on the locator can display a plot, bar graph orany suitable form of display format that is derived from the spectralscan that is shown in FIG. 6a such that the operator is then allowed tomanually select one of the sub-bands, for example, by touching thesub-band of choice on the display screen. In still another embodiment,locator 20 can allow the operator to initially enter informationrelating to the transmitters that are available for automatic selectionof a sub-band that is covered by one of those transmitters, excludingsub-bands that are not available, in a manner that is consistent withthe teachings of U.S. Pat. No. 8,729,901 which is commonly owned withthe present application and hereby incorporated by reference in itsentirety. FIG. 6b illustrates sub-bands that are not available, based onunavailable transmitters, using dashed lines. Conversely, solid linesindicate sub-bands that are available. In the present example, SB-1 andSB-5 through SB-7 are not available. In an embodiment, sub-bands can beexcluded based on regulatory constraints. In this way, the portabledevice itself and the operator are not allowed to make frequencyselections that would violate regulations in a particular jurisdiction.Such frequency restrictions can be predetermined by the manufacturer ona regional basis. In an embodiment, portable device 20 or some othercomponent of the system such as, for example, drill rig 80 can beequipped with a GPS receiver that can establish the location of theinground operation and then look up the local frequency requirements.

Still referring to FIG. 6b , the display screen that is shown can remain“live” at least until the frequency selection process is completed. Thatis, the average noise per sub-band can be monitored and displayed,either alone or weighted by other statistical parameters, in real timefor operator monitoring purposes. In this way, the operator can move thelocator about while observing the average noise in the varioussub-bands. For example, the operator can walk a planned borepath andmonitor the noise along the borepath prior to beginning drilling. Inthis way, a sub-band that is particularly noisy at one or more pointsalong the borepath can be avoided. If the operator so chooses, he or shecan move the locator to a different point, for example, along theborepath and initiate a rescan of the noise across the entire bandwidthby selecting a rescan button 784. As discussed above, the noiseenvironment can be characterized based on reception using one or moreantennas. The operator can change the receiving mode using a button 786.For example, in one receiving mode, the bar chart of FIG. 6b can bepresented based on reception along a single axis such as, for example,the X axis. In another receiving mode, the bar chart can be presentedbased on a vector sum produced from three orthogonal receiving axes.Once the operator changes receiving modes, rescan button 784 caninitiate a new noise scan and present the noise values based on theselected receiving mode. The operator can switch between the variousnoise scanning modes at will. In an embodiment, the noise scan thatforms the basis for the display of FIG. 6b can be a high resolutionscan. In conjunction with performing the noise scan, a number ofoptimized, low noise frequencies can be selected automatically based onthe number of symbol frequencies that is needed. For example, sixteensymbol frequencies and a depth frequency can be selected per sub-band.In an embodiment, during the presentation of live noise on the screen ofFIG. 6b , noise per sub-band can be presented as an average of the noisevalues measured at each of the selected frequencies within eachsub-band. It is noted that selection of rescan button 784 causes a newor updated selection of frequencies within each sub-band. Locator 20 canbe configured to store sets of frequency selections that are associatedwith different measurement positions, for example, in memory 714 of FIG.9. Accordingly, the frequency selections are optimized for eachmeasurement position such that different selections can be used atdifferent times during the operation. The term “optimized” is intendedto mean that the selected frequencies are chosen with the intent ofavoiding interference based on one or more statistical parameters suchas, for example, average noise, standard deviation and peak noise. Thefrequency selection sets can be communicated to the transmitter, forexample, above ground using external communication connection 174 ofFIG. 3. An inground transmitter can be commanded in any suitable mannerto switch to a different set of frequency selections during the ingroundoperation. For example, switching can be commanded based on apredetermined roll sequence of the drill string or by transmission of anelectromagnetic signal from above ground for reception by transmitter130 which is, in this case, configured as a transceiver. Someembodiments can use the drill string as an electrical conductor or caninclude a well-known wire-in-pipe arrangement such that data can betransmitted between the inground transmitter/transceiver and the drillrig. For example, the drill rig can send a command via the drill stringto cause the depth frequency to change.

Attention is now directed to FIG. 10a which is an expanded view ofsub-band 6 from FIG. 6a , generally indicated by the reference number800. For purposes of the present discussion, it will be assumed thatSB-6 is available and has been selected by the operator for use duringthe inground operation. Having selected a sub-band, the frequencies fordepth signal 120 and data signal 122 can be established. In anembodiment, the frequencies can be predetermined, for example, by themanufacturer or based on a previous noise scan, as described above. Inanother embodiment, display 36 can be used to represent the spectralplot of FIG. 10a , in any suitable form, to an operator of the locatorsuch that the operator can make frequency selections. FIG. 10billustrates one embodiment of a screen shot which shows display 36illustrating SB-6. It should be appreciated that the locator can providea zoom function on display 36 that uses Zoom In button 802 and Zoom Outbutton 804 such that the operator can expand the horizontal extents ofthe spectral display to provide for detailed frequency selection.Generally, the operator can select frequencies that correspond to lownoise points on the displayed spectrum. The selections can be rounded toreflect the frequency resolution of the transmitter that is to be used.As discussed above, embodiments of transmitters according to the presentdisclosure can have a frequency resolution of 5 Hz, by way ofnon-limiting example. Twenty-one low noise points are identified on FIG.10a indicated as upticks (a)-(u). In an embodiment using one depthfrequency for depth signal 120 and 16 symbol frequencies, seventeen ofthese 21 frequencies can be utilized. As described above, the depthfrequency can be located at any position within the sub-band,intermingled with the symbol frequencies, at either end of the sub-bandor even in a different sub-band. As one example, the depth frequency canbe selected as the lowest noise point among the identified frequencies,which is frequency (j) in the present example. In still anotherembodiment, the frequencies can be automatically picked or re-picked bylocator 20, for example, responsive to the operator selecting an“Auto-Pick” button 806 on display 36. In one embodiment, processor 710can examine the spectrum of FIG. 6a to identify the lowest noise pointsuntil a suitable number of symbol frequencies is available. In otherembodiments, the processor can perform the selection process based onany suitable method. For example, the lowest noise frequencies can beselected in conjunction with maintaining a minimum separation betweenadjacent frequencies.

Still referring to FIG. 10b , a frequency can be added, for example, bytouching an Add Frequency button 808 and then touching the spectralplot. A frequency can be deleted, for example, by touching a DeleteFrequency button 810 and then touching the frequency to be deleted. Afrequency can be moved, for example, by touching a Move Frequency button812 and then touching and dragging the frequency to be moved. Theselected sub-band can be changed by touching a Change Sub-Band button814. As will be further described immediately hereinafter, frequencyselection is not limited to identification of low noise points but alsocan consider high noise points or areas of the spectral scan.

FIG. 11 is a further expanded view of the spectral region of FIG. 10afrom 24 KHz to 25 KHz, generally indicated by the reference number 820and shown here for purposes of describing further details with respectto frequency selection. In addition to identifying low noise points, asdescribed with regard to FIG. 10a , processor 710 can apply what can bereferred to as a “keep-out region”. The later will exclude anyidentified low noise frequency having a noise peak within a selectedfrequency window 822 that is centered on that low noise frequency. Thenoise can be identified, for example, based on a magnitude that exceedsa threshold 824 based on the average noise value for the sub-band and/orthe noise value associated with the nearby low noise point. In anembodiment, the frequency window can be approximately 60 Hz (+/−30 Hz)in width and the threshold can be 10 dB or more above the associated lownoise point. Based on the use of such a frequency window, frequencies(b) and (e) can be excluded due to the proximity of peaks 826 and 830,respectively. In the event that more frequencies are needed, processor710 can re-examine the spectrum of FIG. 11 for purposes of identifying anew set of frequency candidates.

FIG. 12 is a flow diagram that illustrates an embodiment of a method,generally indicated by the reference number 900, for the operation oflocator 20 in performing spectral scanning and frequency assignment inaccordance with the present disclosure. The method begins at 904 andproceeds to 908 which performs a scan of the full frequency spectrum,for example, from 0 Hz to 45 KHz for the present embodiment, althoughany suitable range can be used for this scan. The scan can be ahigh-resolution scan, for example, utilizing a resolution of 5 Hz, asdiscussed above. In another embodiment, an initial, lower resolutionscan can be utilized such that the resolution is just sufficient toestablish an average noise value for each sub-band. If the sub-bandselection process relies on a lower resolution spectral scan, a highresolution spectral scan can subsequently be performed as part of thefrequency selection procedure, described below. When a widebandtransmitter will be used for the inground operation, a single highresolution scan can be employed for frequency selection purposes. At910, the average noise value per sub-band is determined. At 914, asub-band can be recommended based on the average noise values.Generally, the sub-band having the lowest average noise value can berecommended, although other embodiments can utilize differentrecommendation protocols. For example, the sub-band having the lowestnoise peak value can be recommended. By way of another example, asdiscussed above, more than one sub-band can be recommended. At 918, userinput can be requested on display 36 wherein the user can accept therecommended sub-band or choose a different sub-band. For example, theuser may choose a different sub-band based on an awareness oftransmitters that are available for performing the inground operation.As discussed above, this information can serve as an initial input suchthat method 900 excludes sub-bands that are not covered by the currentlyavailable transmitter(s). Once the sub-band has been selected, themethod proceeds to 920 which determines the sub-band frequencies. In anembodiment, the sub-band frequencies can be predetermined and stored inmemory 714 of the locator or in memory 218 of the transmitter. Inanother embodiment, the sub- band frequencies can be determined by theoperator on-the-fly by presenting the sub-band on display 36, asdiscussed above. In another embodiment, the sub-band frequencies can bedetermined automatically in accordance with the discussions relating toFIGS. 10 and 11 with or without the application of a keep-out window asapplied by step 924. In the instance of the use of a widebandtransmitter for performing the inground operation, steps 910 and 914 arenot required since the entire transmission bandwidth can be availablefor frequency selection without the need to confine the frequencies toany particular sub-band(s) and step 920 can allocate frequencies acrossthe entire transmission bandwidth. Accordingly, transmission frequenciescan be selected automatically across the entire available bandwidthand/or customized by the user based on a high resolution noise scanwithout the need for frequency assignment limitations based onsub-bands. It should be appreciated that an embodiment of a widebandtransmitter can be configured to operate, for example, based on operatorpreference, using sub-bands in the same manner as sub-band limitedtransmitters wherein frequency assignment can be confined to one or moresub-bands, although this is not required. At 928, a determination ismade as to whether a sufficient number of frequencies have beenidentified. If not, operation returns to 920 for identification ofadditional frequencies. If a sufficient number of frequencies have beenidentified, operation proceeds to 930 which recommends frequencies fordepth signal 120 and data signal 122. This latter step may be optionalin a fully automated embodiment. At 934, information can be presented ondisplay 36 for purposes of gathering user input, for example, approvingthe frequency selections or changing the frequency selections. Forinstance, the user may prefer to move the depth frequency to a differentlocation within the sub-band or to an altogether different sub-band. Ofcourse, in a wideband transmitter embodiment, no restrictions need beimposed with respect to limiting frequency selection to a particularband and/or sub-band. At 938, the frequency selections can betransferred to transmitter 130 using external communication arrangement722 of the locator and external communication link 174 (FIG. 3) of thetransmitter. Normal operation can then be entered at 940.

In an embodiment of method 900, the number of frequencies that isselected can be based on the noise environment. For example, if a noisescan, whether sub-band limited or not, shows a low noise environment,relatively more frequencies can be selected. In this case, 32 or moresymbol frequencies can be used instead of 16 symbol frequencies. If thenoise scan shows a high noise environment, relatively fewer symbolfrequencies can be used such as, for example, 4 or 8 symbol frequenciesinstead of 16 frequencies. Generally, the use of relatively fewerfrequencies can aid in avoiding variable noise sources in a highinterference environment. On the other hand, using a higher number ofsymbol frequencies can increase data throughput.

FIG. 13 is a flow diagram illustrating an embodiment of a method foroperation of locator 20 in a normal mode during an inground operation,generally indicated by the reference number 1000. The method begins at1004 and proceeds simultaneously along a depth determination branch 1010and a data recovery branch 1012. Depth branch 1012 receives depth signal120 at 1020 and then determines the depth of the transmitter at 1024.Because the depth signal is transmitted on a dedicated frequency, thedepth signal is receivable on an essentially continuous basis throughoutthe inground operation. Accordingly, steps 1020 and 1024 repeat in aloop fashion throughout the normal operation mode of the locator. Asdescribed above, step 1024 can utilize the depth signal to determine thedepth of the transmitter based on the dipole equations. In anembodiment, part of the depth determination can include compensation forthe distance of the locator above the surface of the ground. Datarecovery branch 1012 begins at 1030 with reception of data signal 122 inthe form of a symbol stream that can be made up of multi-bit symbols. At1034, the symbol stream can be temporarily stored for decoding, forexample, in memory 714 (FIG. 9). At 1038, processor 710 decodes thesymbol stream. In this regard, one of the symbols can be used as asynchronization symbol that can identify the start of a packetstructure. In an embodiment that uses a 4 bit symbol (i.e., 16 symbolfrequencies), a seventeenth symbol frequency can be added for purposesof representing a synchronization symbol in the symbol stream. Onesuitable packet structure, by way of non-limiting example, can berepresented by a series of 4-bit variables as S, P1, R1, P2, R2, BT1,BT2, R3 wherein S has a fixed value that corresponds to the sync symbol,P1 is a variable representing the first four bits (0-3) of a pitchvalue, R1 is a first roll variable characterizing the roll orientation,P2 is bits 4-7 of the pitch value, BT1 is a first four bits (0-3) ofbattery and temperature data, BT2 is bits 4-7 of battery and temperaturedata, and R3 is a third roll variable. In this regard, it should beappreciated that the pitch value is accumulated based on two differentvariables corresponding to two symbols in the symbol stream that areseparated by another symbol. That is, the four bits of P2 can beappended to the four bits of P1 to represent a complete pitch value.Still further bits can be appended based on another pitch variable, ifdesired. Similarly, 8 bits of battery and temperature data can beassembled based on two successive variables BT1 and BT2. At 1040, a datastream can be reassembled based on the decoded symbol stream toreconstruct the original data that was the basis for the symbol streamin transmitter 130. At 1044, processor 710 recovers parameters from thedata stream. These parameters can represent orientation parameters suchas pitch and roll, temperature, pressure, battery voltage and current,and any other data that is of interest. At 1048, processor 710 respondsto the recovered parameters in any suitable manner such as, for example,by driving display 36 to indicate pitch and roll, battery status,temperature and pressure and/or as inputs for other processes such as,for example, providing warnings when thresholds relating to temperatureand pressure have been violated. Operation then returns to step 1030.

Still considering the operation of transmitter 130 in the normal mode,it should be appreciated that transmission power can be allocatedunevenly between the various frequencies that are transmitted such as,for example, those that are shown in Table 3. In one embodiment, eachfrequency can be allocated an equal amount of transmission power. Inanother embodiment, transmission power can be allocated non-uniformlyamong the frequencies. For example, one or more frequencies can beassigned a higher transmission power than another group of frequencies.In still another embodiment, each frequency can be assigned a differenttransmission power. Such power allocation can be performed in anysuitable manner. For example, in an embodiment, portable device 20 ofFIGS. 1 and 9 can be configured to monitor the average signal strengthassociated with each frequency as each frequency is received duringnormal operation. Transmission power can then be reallocated on-the-flyamong the frequencies based on a running average signal strength. Forexample, a sudden increase in signal strength of a given frequency willgenerally be attributable to interference such that additional power canbe allocated to that frequency. In some embodiments, low noisefrequencies can be allocated relatively lower transmission powers whilehigher noise frequencies can be allocated relatively higher transmissionpowers. The reallocated transmission power values can be transferred totransmitter 130 in any suitable manner. For example, portable device 20can transmit the reallocated power scheme to drill rig 80 via telemetrysignal 44. The drill rig can then transfer the new power scheme totransmitter 130 via the drill string by using the latter as anelectrical conductor. In another embodiment, portable device 20 can beconfigured with an additional antenna 712 (FIG. 1) such as a dipoleantenna for transmitting a signal for direct reception by transmitter130. Modulation of this signal can be decoded by transmitter 130 torecover the new power scheme.

Referring again to FIG. 10b , power allocation among selectedfrequencies can also be performed during the frequency selection processor mode, prior to entering normal operation. For example, powerallocation can be based on a noise value that is associated with eachselected frequency, as shown in FIGS. 10 and 10 a. Although the lownoise frequencies identified in FIGS. 10 and 10 a appear to exhibitrelatively equal noise values for illustrative purposes, this may notnecessarily be the case. If there is significant variation among thenoise values for the lowest noise frequencies that are identified,transmission power can allocated in a higher proportion to thosefrequencies exhibiting relatively higher noise values. Conversely,transmission power allocated to a very low noise frequency can berelatively low to allow for additional power allocation to one or moreother frequencies. Transmission power can also be allocated in a mannerthat is consistent with the application of a keep-out window, asdescribed above. For example, if a particular frequency is selected suchthat a powerline harmonic or other noise anomaly falls within a keep-outwidow for that particular frequency, additional power can be allocatedto the particular frequency. It should be appreciated that in anyembodiment that uses allocated transmission power that can vary fromfrequency to frequency, such allocation can be performed based onoverall power consumption, particularly when transmitter 130 is batterypowered. In this way, the overall power consumption can be reduced or atarget overall power consumption can be maintained.

Turning now to FIG. 14, in another embodiment, system 10 can beconfigured for depth determinations based on data signal 122 such thattransmission of the depth signal is not necessary. It is noted thatreceiver 20 of FIG. 9 can be configured to include any of the describeddetectors described hereinafter. FIG. 14 is a block diagram of anembodiment of a depth detector, generally indicated by the referencenumber 1100 that forms part of locator 20. In this embodiment, a seriesof n band-pass filters BP_(a)-BP_(n) are shown with each band-passfilter being wide enough to allow symbols of each frequency to passtherethrough. In an embodiment, a band-pass filter is provided for everysymbol frequency a-n. The filter outputs are provided to respective onesof amplitude detectors 1108 a-1108 n each of which provides an output toa peak amplitude router 1110 which monitors the outputs of the amplitudedetectors to determine which amplitude detector is providing the highestoutput value. During operation, peak amplitude router 1110 transfers thehighest detected amplitude output to a low-pass filter 1120. An output1122 of the low-pass filter is proportional to the depth of thetransmitter for purposes of depth calculation.

Referring to FIG. 15, another embodiment of a depth detector isgenerally indicated by the reference number 1100′. In this embodiment, nmultipliers M₁-M_(n) multiply each frequency or tone by a differentfrequency that is offset by a constant amount from one frequency to thenext. The resulting signal for each frequency or tone is comprised ofthe sum of the two frequencies and the difference of the two frequencieswhich is equal to the offset. For n multipliers each with the correctmultiplier frequency, the outputs can be added by an adder 1140resulting in a phase continuous signal 1144 at the offset frequency. Thephase continuous signal is provided to a narrow band-pass filter 1148.An output 1150 of band-pass filter 1148 is then proportional to thedepth of the transmitter.

Referring to FIG. 16, still another embodiment of a depth detector isgenerally indicated by the reference number 1160. In this embodiment, aninput 1162 receives input symbol stream 122 subject to an errorcorrection code such as, for example, a Reed Solomon (RS) code. Thelatter can be applied by an EC (Error Correction) Encoder 1164 of FIG. 3via processor 200. FIG. 17 illustrates a Reed Solomon codeword as partof the symbol stream including a block of data symbols 1166 startingwith a sync symbol to which is appended a block of RS correction orparity data 1168. It is noted that data symbols and Reed Solomon symbolscan be transmitted in a variety of different ways. For example, the RSsymbols can be appended as a block, as illustrated. In anotherembodiment, the RS symbols can be interspersed among the data symbols.The input symbol stream is received by a demodulator 1170 which recoversthe data stream subject to potential transmission errors. It is notedthat the RS code is capable of correcting up to a certain number ofsymbols per RS codeword, based on parameters that are well-known. In thepresent embodiment, up to two symbols can be corrected. The recoveredsymbol stream is transferred to an accumulator 1172 which can buffer therecovered data for purposes of identifying the RS codewords as part ofthe data stream. As part of the data, a received amplitude can beassociated with each data symbol. The start of a codeword, for example,can be identified based on the presence of a synchronization symbol.Once a codeword is available, the codeword can be transferred to anError Correction (EC) Decoder 1174. The EC decoder then decodes thecodeword and performs corrections up to the limit of the correctioncapability of the code. Each corrected symbol is identified as suchduring this process. At 1176, an amplitude estimator associates correctamplitudes with the symbols that were corrected by the RS code. In thisregard, it should be appreciated that a symbol that was incorrectlyidentified by demodulator 1170 is misassociated in terms of symbolfrequency. Estimator 1176 corrects this misassociation. The correctedsymbol frequency amplitudes are transferred to low pass filter 1120. Anoutput 1122 of the low-pass filter is proportional to the depth of thetransmitter for purposes of depth calculation. In an embodiment, thedepth determination can be based on a number of amplitude values fromone or more symbol frequencies.

Attention is now directed to FIG. 17 which is a flow diagramillustrating an embodiment of a method, generally indicated by thereference number 1180 for depth determination based on symbolfrequencies and without the need to transmit a depth signal. The methodbegins at 1182 and proceeds to 1184 which encodes and transmits symbolstream 122 from an embodiment of a transmitter in accordance with thepresent disclosure, subject to an error correction code such as, forexample, a Reed Solomon code. At 1186, the symbol stream is received byreceiver 20 (FIG. 9). At 1188, the received symbol stream is bufferedand the current RS codeword is recovered. At 1190, the codeword isdecoded. At 1192, if it is determined that the current codeword is notdecodable, since the number of symbols in error exceeds the correctioncapability of the code, operation is routed to 1186 to receive the nextcodeword such that the current codeword is ignored at least in terms ofcontributing to depth determinations. On the other hand, if the codewordis decodable, operation is routed to 1194. At 1194, if no symbols werecorrected, operation proceeds to 1196 which utilizes depth detector 1100of FIG. 14 to determine the depth of the transmitter. Operation thenreturns to 1186 to receive the next symbol. If one or more symbols werecorrected, step 1194 routes operation to 1198 which utilizes errorcorrection based depth detector 1160 of FIG. 16 to determine the depthof the transmitter.

Referring to FIG. 18, an embodiment of antenna 26 of FIG. 1 is shown ina further enlarged diagrammatic view for purposes of illustratingdetails of its structure. Antenna 26 includes first and second ferriterod antennas 1200 a and 1200 b, each supporting a respective antennacoil 1204 a and 1204 b representing antennas Y and Z of FIG. 9. Theferrite rod antennas can be supported on opposite major surfaces of aprinted circuit board (PCB) 1210. The latter defines an antenna coil1214 (partially shown) that serves as antenna X in FIG. 9. As shown inthe latter figure, the PCB is vertically oriented at least generallyalong the axis of the locator. It is noted, however, that the antennamay be oriented in any suitable manner within locator 20 and is notlimited to the orientation that is shown.

In view of the foregoing, it is submitted that the system, associatedapparatus and methods of the present disclosure sweep aside thelimitations of the prior art with respect to attempt to the prior arthit or miss proposition of identifying a “magic” frequency. In contrast,the use of a multi-bit symbol stream structure, which assigns adifferent frequency to each symbol, can serve to enhance the datathroughput rate and/or limit the transmitted spectral content to provideenhanced noise immunity. Further, the present disclosure provides forfrequency customization which affirmatively avoids noise interference ina given region. This frequency customization can even be different atdifferent times for the same region. Further, the present disclosurebrings to light capabilities with respect to avoiding power lineharmonic frequencies that are submitted to be heretofore unknown atleast with respect to horizontal directional drilling for purposes ofutility installation. In particular, symbol frequencies that aretransmitted in the context of the present disclosure are modulated in amanner that places essentially no spectral content at adjacent powerline harmonics. Prior art techniques, in contrast, often utilize aspectral modulation envelope that overlaps a number of powerlineharmonics, thereby placing signal content on top of the powerlineharmonics. On the receiver side, the overlapped powerline harmonics arethen necessarily picked up along with the modulated signal. The presentdisclosure, in contrast, utilizes symbol frequency reception, based onthe transmission technique that has been brought to light, whichsuppresses adjacent powerline harmonics. These capabilities arise atleast on the basis of the transmission of a multi-bit symbol stream in amanner that provides for precision positioning of symbol frequencieswhile limiting and positioning the spectral bandwidth that is associatedwith each symbol of the symbol stream in a heretofore unseen mannerrelative to powerline harmonics.

In another embodiment, system 10 can utilize what may be referred to as“silent sync” or “null sync.” In this embodiment, instead oftransmitting a sync symbol, a symbol interval is left blank, null orsilent. This symbol interval may be referred to hereinafter as a nullsymbol. FIG. 19 illustrates another example output 390′ of symbolchannel PWM generator 352 (FIG. 4) in which the symbol stream, indicatedas symbols 392 b-392 f, is preceded by a null symbol 1300 having thesame time duration as one of the symbols in the symbol stream, althoughany suitable duration can be used. It should be appreciated that silentsync can be used with forms of modulation including, but not limited toBinary Phase Shift Keying (BPSK), Differential Binary Phase Shift Keying(DBPSK), Manchester encoding, Quadrature Phase Shift Keying (QPSK),M(ary)PSK and so on.

FIG. 20 illustrates an embodiment of a transmitted data stream,generally indicated by the reference number 1240, illustrating furtherdetails with respect to silent sync. For purposes of the presentexample, it is assumed that a form of modulation such as BPSK is used,although the present descriptions enjoy broad applicability with respectto modulation type. A frame interval F includes an overall duration thatcan correspond to a packet or frame of data. Frame interval F is made upof a sync portion S and a data portion D, comprising information thatcan be modulated based on one or more carrier frequencies. Sync portionS can be made up of a null symbol, NS, or null bit and a reference bit,RB. In the example of BPSK, a single carrier frequency can be utilized.

Referring to FIG. 21, a portion of a received signal, corresponding tothe transmitted data stream of FIG. 20, is generally indicated by thereference number 1300. The received signal, subject to noise, iscentered on 0 volts and is plotted against time on the horizontal axis.Sync portion S is shown along with part of data portion D. FIG. 22illustrates a waveform 1320 that is the squared value of the waveform ofFIG. 21. Although squaring is not required, based on squaring thewaveform, null symbol NS is seen to exhibit a dramatically lowermagnitude in FIG. 22 than the peak amplitudes that are present inreference bit, RB, and data portion D of the waveform such that the nullsymbol can readily be identified, for example, by storing the receivedsignal in a buffer. It is noted that this technique can be effectiveeven in environments with relatively high levels of interference and/orat depths of the boring tool that significantly decrease thesignal-to-noise ratio. In high noise environments, ensemble averagingcan be applied wherein multiple frames of data can be stored in buffermemory and then added for purposes of recovering synchronization.Ensemble averaging is described in detail, for example, in U.S. patentapplication Ser. No. 14/208,470 which is commonly owned with the presentapplication and incorporated herein by reference.

Attention is now directed to FIG. 23 which illustrates an embodiment ofa method for operating the system of the present application usingsilent sync, generally indicated by the reference number 1400. Themethod starts at 1402 and proceeds to 1404 which transmits a data streamincluding a null sync portion or symbol. At 1406, the data stream isreceived. At 1408, the data stream can be squared. For relatively lownoise environments, it should be appreciated that squaring the datastream may not be necessary. At 1408, the data stream is decoded,premised on an identification of the null sync symbol. At 1410, anattempt to decode the squared data stream is made including anidentification of the null sync symbol. At 1412, if decoding wassuccessful, operation proceeds to 1414 such that the recovered data isused. On the other hand, if the decoding attempt was unsuccessful,operation can proceed to 1416 such that ensemble averaging is applied.At 1418, data is recovered based on the ensemble averaging. Therecovered data is then used at 1414.

It is well-known that the accuracy of a system for electronicallymeasuring the depth of a boring tool while underground can be impairedas a result of skin effect that results from the conductivity of theearth. Without compensation, error can be introduced that can cause theboring tool to appear to be at a depth that is less than its actualdepth. Techniques to compensate for skin effect error, as limited toinfluencing depth readings of the boring tool, are described in detailin U.S. Pat. No. 6,285,190 (hereinafter, the '190 patent) which iscommonly owned with the present application and incorporated herein byreference. Applicants further recognize herein that a similar phenomenonat the surface of the ground, hereinafter referred to as “surfaceeffect,” can impair the accuracy of such a system while it is operatedabove ground, for example in connection with a customer demonstration ortesting of the system. That is, with the boring tool and a portabledevice separated at the surface of the ground by a known distance, asignal-strength-based distance between the two, as determined by theportable device, can vary dramatically from the known distance absentsurface effect compensation. While one might assume that the solution isto perform skin effect calibration at the surface of the ground tocompensate for the surface effect, the '190 patent recognizes that skineffect calibration at the surface of the ground is problematic duringthe actual drilling operation, since significant depth error can beencountered as compared to the depth error that is seen in response to abelow ground skin effect calibration procedure. Based on the unresolveddifficulties that are presented by the foregoing, Applicants bring tolight a multi-mode arrangement which operates in an above ground mode toprovide for enhancing signal-strength-based accuracy for above groundreadings relating to surface effect and a below ground mode that doesnot apply the same surface effect compensation as the above ground modesuch that below ground performance is not compromised with respect todepth reading accuracy. Accordingly, surface effect compensation in theabove ground mode is different than the compensation, if any, that isapplied in the below ground mode. In some embodiments, the below groundmode can apply skin effect compensation. In other embodiments, the belowground mode can operate without skin effect compensation.

Referring to FIG. 24, a plot is generally indicated by the referencenumber 1500 and illustrates surface effect error plotted on the verticalaxis versus horizontal distance, on an X axis, between a portable devicesuch as, for example, portable device 20 of FIG. 1 and a boring tool, orother transmitter that transmits an electromagnetic signal such as, forexample, depth signal 120 of FIG. 2 or a modulated signal from whichdepth can be determined based on signal strength. Surface effect error1504 can be characterized as:

SEE=kx^(3 tm (EQN) 2)

Where x is the separation or distance between the boring tool and theportable device, SEE is the Surface Effect Error and k is a constant fora particular measurement location. The constant k can vary by locationas a result of active and/or passive interference in the region, soilconditions and other factors. It is noted that the SEE can be expressedin terms of any suitable function and is not limited to a cubicfunction. A calibration procedure based on Equation (2) can be performedby placing the portable device at a known or measured distance, d_(m),from the boring tool at the surface of the ground. Applicant hasidentified that a suitable value of d_(m) for performing the calibrationprocedure can be 50 feet; however, calibration can be performed withother values of d_(m) so long as (1) the value of d_(m) is not so smallthat the measured SEE at a particular measurement location is notappreciable, and (2) the value of d_(m) is not so large that it isbeyond the range of the measuring device. With this physical arrangementin place, the portable device determines a distance, d_(ss), to theboring tool based on the signal strength of depth signal 120. Thesurface effect error value for measured distance d_(m) is shown in FIG.24 as an offset from the X axis at d_(m) and can be determined based on:

SEE=d _(m) −d _(ss)   (EQN 3)

It should be appreciated that the value of SEE will generally be greaterthan zero, since the surface effect typically causes d_(ss) to be lessthan d_(m). Equation 2 can now be solved for the value of constant kusing the value of SEE determined from Equation 3 and the measured valueof d_(m). Subsequently, compensation for surface effect can be appliedbased on Equation (2) using the determined value of k whenever the aboveground range (AGR) between the boring tool and portable device is beingdetermined based on signal strength. In particular, a compensated aboveground range, AGR_(COMP), value can be determined for any measuredsignal strength value d_(ss) based on:

AGR_(COMP) =d _(ss)(1+kd _(ss) ²)   (EQN 4)

Based on the foregoing, portable device 20 can be operated in an AboveGround Range Test (AGRT) mode that applies surface effect compensationto any signal-strength-based distance or range that is determined by theportable device. When the boring tool is below ground, portable device20 can be operated in a normal mode that can be configured to determinedepth of the boring tool based on signal strength without applyingsurface effect compensation or by applying skin effect compensation thatis different than the manner in which the surface effect compensation isapplied such that the depth of the boring tool can be established with ahigher degree of accuracy. It is noted that surface effect compensationparameters and skin effect compensation parameters, if used, can bestored in memory 714 of the portable device as seen in FIG. 9.

FIG. 25 is a diagrammatic illustration of an embodiment of theappearance of display 36 (FIG. 1) of portable device 20, operating inthe AGRT mode and generally indicated by the reference number 1600.Boring tool 90 is indicated in a laterally spaced apart relationshipfrom portable device 20 with the surface effect compensated offset valueindicated at 1610 as 10 feet. It should be appreciated that the graphicsshown in the screen shot of FIG. 26 clearly indicate to the user thatdistance to the side of the locator is being shown, as opposed to depthbeneath the locator. The AGRT mode can be entered, for example, based onan operator selection of the AGRT mode that is available on acalibration screen. Otherwise, the portable device operates in thenormal mode.

FIG. 26 is a diagrammatic illustration of an embodiment of theappearance of display 36 (FIG. 1) of portable device 20, operating inthe normal mode, generally indicated by the reference number 1630. Theportable device is seen in both an overhead view (upper left) and anelevational view (upper right). Boring tool 90 and the portable deviceare shown in relation to a forward locate point 1634. For additionalinformation regarding the forward locate point see, for example, U.S.Pat. No. 6,496,008 which is commonly owned with the present applicationand incorporated herein by reference. A lateral distance 1638 from theboring tool to a position directly below the forward locate point isshown as 6 feet, 0 inches. A predicted depth of the boring tool 1640 atthe boring tool, once the forward locate point is reached, is shown as 8feet, 7 inches and a height above ground of the portable device 1644 isshown as 2 feet, 0 inches.

FIG. 27 is a flow diagram that illustrates an embodiment of a method,generally indicated by the reference number 1700, for the operation ofportable device 20 in a dual mode configuration including an AGRT modeand a normal mode. The method begins at 1704 and proceeds to 1708 whichmonitors for a selection by the operator to perform a calibration forthe AGRT mode. If AGRT calibration is not selected, operation proceedsto normal operation at 1710. On the other hand, if AGRT calibration isselected, operation moves to 1712 which initiates a calibrationprocedure, for example, as described above in connection with FIG. 25.At 1714, a determination is made as to whether the AGRT calibration wassuccessful. The determination as to whether or not the AGRT calibrationwas successful can be based on various factors. For example, thedetermined value for constant k can be compared to an acceptable range.As another example, the value of k can be determined at differentdistances between the portable device and the boring tool and thencompared. From the perspective of determining the value of k, based onEquation (20), one would expect the magnitude of the surface effecterror to increase with increasing range between the portable device andthe boring tool. If the calibration was not successful, step 1712 isreentered. If the calibration was successful, operation proceeds to 1718which monitors for operator selection of the AGRT mode. If the operatordoes not select the AGRT mode, normal operation is entered at 1710. Ifthe operator does select the AGRT mode, step 1724 allows the operator toreturn to 1712 for another AGRT calibration. If the additionalcalibration is not selected, the AGRT mode is entered at 1728. Step 1730tests for whether or not the operator wishes to exit the AGRT mode, forexample, based on actuating trigger 48 (FIG. 1). If the operator choosesto exit, normal operation 1710 is entered. Otherwise, operation ismaintained in the AGRT mode. It should be appreciated that normal mode1710 allows the operator to select calibration which can cause operationto return to 1724.

Attention is now directed to FIG. 28 which is a block diagram of anembodiment of a receiver section, generally indicated by the referencenumber 2000 that can be implemented as part of device 20 of FIG. 9 forpurposes of receiving and processing data signal 122. Receiver section2000 includes a slicer for receiving the data signal in slices 2004 suchthat more than one slice is associated with the period of one symbol inthe symbol stream that forms data signal 122. Each time slice may bereferred to as a spectrogram time slice (STS). In the presentembodiment, five time slices are received for each symbol period,although any suitable number of slices may be received. Accordingly,each symbol can be considered as being oversampled. Each STScharacterizes the signal strength at all of the symbol frequencies thatare in use, even though only one symbol frequency is active at any giventime (see, for example, transmitter output 390 of FIG. 4). The activesymbol frequency will generally be significantly higher in amplitudethan the remaining frequencies, dependent upon the current noiseenvironment. In the present example, 16 symbol frequencies 2008 areutilized, although the present description is applicable to any numberof symbol frequencies. At 2010, for a current slice, an average signallevel is determined for the combination of symbol frequencies that is inuse such that the measured signal level at each symbol frequencycontributes to the determined average signal level. The average signallevel is output at 2012. A low pass filter 2014 applies filtering to theaverage signal level. Any suitable form of low pass filter such as, forexample, a low pass Butterworth filter can be utilized. The low passfilter is configured with a time constant that is long compared to thetime period of a symbol such that contributions from instantaneous noiseevents or pulses, represented as part of the average signal level ofeach time slice, are effectively averaged out. In this way, output 2018of the filter tracks or follows the ambient noise level based oncontributions of signal strengths from inactive symbol frequencies foreach slice in combination with a contribution from an active symbolfrequency. A threshold offset section 2020 specifies an incrementalvalue to be added to the average signal level. The value that is added,for example, can correspond to some percentage of the average signallevel such as, for example, a fixed amount. An adder 2022, which formspart of a threshold detector 2024, adds the value to output 2018 of thelow pass filter to output a threshold at 2028. Average signal level 2012is also received by a comparator 2030, which forms another part ofthreshold detector 2024. The comparator compares threshold 2028 toaverage value 2012 and drives a time slice switch 2040 responsive tothis comparison. A spectrogram buffer 2044 is configured to selectivelyreceive time slices from the time slice switch. The spectrogram buffercan include a length that provides for storing all of the time slicesSTS₁ through STS_(n), that are associated with a complete packet. By wayof non-limiting example, if a packet includes 20 symbol positions, someof which can be associated with error correction, and each symbol periodis sliced five times, the spectrogram buffer is configured to store 100time slices such that n=100. If comparator 2030 determines that currentaverage value 2012 for a current time slice is greater than threshold2028, time slice switch 2040 is set to the T (True) position such thatthe current time slice is not routed into one of the STS positions inspectrogram buffer 2044. As will be seen, the appropriate pointer STSposition, in this case, is filled with zeros. On the other hand, ifcomparator 2030 determines that current average value 2012 is notgreater than threshold 2028, time slice switch 2040 is set to the F(False) position such that the current time slice is routed into theappropriate one of the STS positions in the spectrogram buffer. Based onthe foregoing and as shown by an inset plot 2050, it should beappreciated that average signal level 2012 will generally fall belowthreshold 2028, except during noise bursts 2054. Thus, slices that occurduring noise bursts 2054 are blocked from contributing to the packetthat is within the length of spectrogram buffer 2044.

Attention is now directed to FIG. 29 which illustrates an embodiment ofa method, generally indicated by the reference number 2100, for loadingspectrogram time slices into spectrogram buffer 2044 of FIG. 28. Themethod begins at 2104 and proceeds to 2108 which receives data signal122. At 2110, spectrogram buffer 2044 is cleared. At 2114, a spectrogrambuffer pointer is reset to indicate a first one of the STS positions inthe buffer. At 2118, the current slice is demodulated to measure thesignal level at the frequency of each symbol in the slice. In thepresent example, 16 symbol frequencies are being employed such that 16signal levels are determined. At 2120, an average signal level isdetermined based on the measured signal levels. The average signal levelis transferred to a threshold set step 2124 (see item 2020 in FIG. 28),a low pass filter 2128 (see item 2014 in FIG. 28) and a comparison step2130 (see item 2024 in FIG. 28). Threshold set 2124 adds an incrementalvalue to the average signal level, as discussed above, and provides thisvalue to an adder step 2134 (see item 2022 in FIG. 28). LP filter 2128applies low pass filtering to the average signal level, as discussedabove, and provides a filtered output to adder 2134. The latter adds theincremental value to the filtered output to provide a threshold tocomparison step 2130. Comparison step 2130 compares the threshold to thecurrent average value of the current slice. If the current average valueis greater than the threshold, operation is routed to 2136 which loadsthe current STS position (pointed to by the STS pointer) with zeroes. Inthis way, instantaneous noise anomalies are not allowed toinappropriately increase the average value of a particular slice whichcan lead to an increase in decoding errors. On the other hand, if thecurrent average value is less than the threshold, the current STSposition is loaded at 2138 with the measured levels at the symbolfrequencies (16 measured levels in the present example). At 2140, theSTS pointer value is tested to determine whether spectrogram buffer 2044is full. If not, the STS pointer is incremented at 2144. If thespectrogram buffer is full, operation proceeds to 2148 which decodes thepacket. Details of an embodiment of the packet decoding process will bedescribed hereinafter. For the moment, it is noted that, while thelength of the spectrogram buffer is matched to the length of the packet,the actual starting position of the packet within the spectrogram bufferis unknown at this juncture for decoding purposes. Subsequent to 2148,operation returns to 2114 to reset the STS pointer in preparation forthe next packet.

FIG. 30 diagrammatically illustrates the contents of spectrogram buffer2044 resulting from loading the buffer according to method 2100 of FIG.29, as well as additional details relating to this content. In thepresent example, a packet structure comprising 20 symbols is utilizedwherein each symbol is represented by 5 spectrogram time slices in thebuffer. As discussed above, packets are received in what is essentiallyan asynchronous manner. The time period of the overall packet structureand each of its symbols is assured based on sufficient stability of theclocking on both the transmission and reception ends of the process.Data is typically loaded or streamed into the length of the spectrogrambuffer beginning at an arbitrary slice of some arbitrary symbol within acurrent packet and extending into an initial portion of the next packet.The time slices for a given symbol can be consecutive within thespectrogram buffer or can wrap around the end of the spectrogram buffer.Accordingly, the packet structure can begin at an arbitrary packet startposition somewhere in the spectrogram buffer. In the present example, aconcluding portion of a first packet, designated as P1, is loaded intoSTS positions 1-20 and an initial portion of a second packet, designatedas P2, is loaded into STS positions 21-100 to fill the spectrogrambuffer. A packet start position 2160 is at STS 21. Thus, P2 includessymbols S1-S16 and P2 includes symbols S17-S20 comprising data from twoconsecutive, but distinct packets. In order to expedite synchronizationof the process being described, transmitter 130 (FIG. 1) can be heldmotionless such that packet content is essentially constant from onepacket to the next and the contents of the spectrogram buffer areessentially representative of a single packet. Data signal 122, forpurposes of the present example, is encoded with a forward errorcorrection code. While any suitable form of forward error correctioncode can be used, the present embodiment employs a Reed Solomon errorcorrection code such that at least one block of error correction dataaccompanies measured and other data of interest. Thus, packet data 2304,as part of the packet structure shown in FIG. 30, can representorientation parameters such as pitch and roll, as well as parametersthat are indicative of the operational status of transmitter 130(FIG. 1) such as, for example, temperature, pressure and battery status.The packet data is followed by an overall block of error correction datathat is made up of a first portion of error correction data RS 1,indicated by the reference number 2308, and a second portion of errorcorrection data RS 2, indicated by the reference number 2310 such thatthe overall block of correction data carries over from the end of thespectrogram buffer back to its beginning. Through the use of the ReedSolomon error correction code, a particular number of errors can becorrected for each packet, dependent upon the size of the block of errorcorrection data. For purposes of the present example, the errorcorrection code is capable of two corrections per packet although anysuitable correction power can be utilized. It should be appreciated thatthe illustrated packet structure is not intended as being limiting andthat any suitable packet structure can be utilized while remainingwithin the scope of the teachings that have been brought to lightherein.

Attention is now directed to FIG. 31 in conjunction with FIG. 30. Theformer is a flow diagram illustrating an embodiment of a decode process,generally indicated by the reference number 2400, that can be employedas decode step 2148 of FIG. 29. The method begins at 2404 and proceedsto 2408 which sets the STS pointer to 1. The latter can be incrementedor set to point to any STS index value (1-100) shown in FIG. 30. At2410, decoding is applied under the assumption that the STS pointer ispointing to an STS slice that represents a slice within the first symbolof the overall packet structure. In particular, the data that isutilized by the Reed Solomon decode process starts with the data at STS1 and utilizes every fifth subsequent slice within the spectrogrambuffer. That is, for the initial decode, slices 1, 6, 11, 16, 21, 26,31, 36, 41, 46, 51, 56, 61, 66, 71, 76, 81, 86, 91 and 96 are used. Inthis way, one slice from each of the 20 symbols in the overall packetstructure contributes to an attempt to correctly decode what may bereferred to hereinafter as a slice packet. Each slice of a slice packetis taken from the same slice position of each symbol. At 2414, thenumber of Reed Solomon error corrections for the decode attempt iscompared to the maximum correction power of the correction code, whichin the present example, is assumed to be 2. If the number of correctionsis less than or equal to 2, operation proceeds to 2418 which stores thedecoded data for the slice packet. Operation is then routed to 2420. Onthe other hand, if 2414 determines that the number of corrections isgreater than 2, operation is routed directly to 2420 which determineswhether the STS pointer is set to the last STS index value in thespectrogram buffer. If not, step 2424 increments the STS pointer by 1and returns operation to RS Decode step 2410. According, by looping backto step 2410, an RS decode attempt is made starting from each STS indexposition in the spectrogram buffer. Each time that a successful decodeis achieved, step 2418 stores the decoded data. Step 2414 stores thenumber of corrections that were made in association with eachsuccessfully decoded slice packet and can designate slice packets thatwere not successfully decoded as undecodable. Table 4, by way ofnon-limiting example, indicates the results of the decode attempts forslice packets 1-100. Successful decodes were achieved for slice packets21-25, while all the remaining slice packets were found to beundecodable.

TABLE 4 START SLICE NUMBER OF CORRECTED ERRORS 1-20 ALL SLICE PACKETSUNDECODABLE 21 1 22 0 23 0 24 0 25 2 26-100 ALL SLICE PACKETSUNDECODABLE

With reference to FIG. 30, it should be appreciated that slices 21-25correspond to the set of slices that is associated with the first symbol(S1) of the packet. On this basis, the center of the initial symbol ofthe packet has been identified as corresponding to slice 23. While allfive slice packets starting with slices 21-25 of symbol S1 might bedecoded with no errors, it should be appreciated that errors can beintroduced, for example, by noise. In such a case, one or more of theslices of one of these slice packets may have been loaded with zerosduring method 2100 of FIG. 29.

Returning to the discussion of FIG. 31, once step 2420 detects the lastSTS index, operation proceeds to 2428 which finds the best STS index orslice packet based on the number of corrected errors in Table 4. In thepresent example, STS 23 is identified as corresponding to the centerposition of the starting symbol of the packet. In an embodiment, thedata that is decoded for the slice packet associated with STS 23 can beconsidered as the best data for the packet since each slice of the slicepacket originates from a center position of each symbol. With regard tothe correction process, the likelihood of decoding erroneous data (i.e.,incorrectly correcting the data) when using the described process isvery low. Of course, the probably of incorrectly decoding or recoveringdata decreases in proportion to the correction power of the forwarderror correction code. Most often, the process either successfullydecodes a slice packet, recovering the correct data, or identifies theslice packet as undecodable. Undecodable slice packets can result, forexample, from noise bursts such that a significant number of slices inthe slice packet are loaded with zeros. At 2430, the method checkswhether the run length for the current packet exceeds a run lengththreshold. The latter can be set to the number of slices per symbol.Under high signal to noise ratio conditions (low ambient noise), thenumber of successful slice packet decodes should correspond to but notexceed the number of slices per symbol. In the present example, if morethan five slice packets are successfully decoded for a packet, there isa high probability that one of slice packets was incorrectly decoded,since that slice packet inherently starts in a symbol that is not thefirst symbol of the packet. If the run length exceeds the run lengththreshold, the packet is identified as undecodable at 2436. If the runlength does not exceed the run length threshold, the decoded data isstored and/or transferred for use by other processes at 2438. The datacan include, by way of non-limiting example, pitch, roll, batterystatus, temperature and pressure. At 2440, the STS pointer is set to astarting position for decoding spectrogram buffer 2044. In oneembodiment based on the example of FIG. 30 and Table 4, the startingposition can be STS 21, the first slice of the packet). In anotherembodiment, the starting position can be STS 23, the center slice of thefirst symbol of the packet. In still another embodiment, once a givensymbol in the spectrogram buffer has been identified as the first symbolof the packet, subsequent decoding can, at least initially, be limitedto decoding the set of slice packets that is associated with the givensymbol.

Referring to FIGS. 9 and 30, in an embodiment, receiver 20 can beconfigured to receive data signal 122 as characterized by an in-phasecomponent, I, and a quadrature component, Q. On this basis, spectrogrambuffer 2044 can be configured for storing two sets of magnitudes for thesymbol frequencies of each STS. One set of magnitudes represents thein-phase components and the other set of magnitudes represents thequadrature components. Stated in another way, the magnitudes associatedwith each time slice are stored in a complex format. Although notrequired, Applicants have discovered a process that is beneficial forpurposes of determining the average magnitude of each symbol frequency.First, the in-phase, I component, subject to noise, for each symbolfrequency in a slice is averaged separately from an average that isdetermined for the quadrature, Q component, subject to noise, of eachsymbol frequency in the slice. Second, the two averages are added.Third, the square root of the sum is taken. This process can beexpressed by equation (5), as follows:

$\begin{matrix}{{{Symbol}\mspace{14mu} {Freq}\mspace{14mu} {Av}\mspace{14mu} {Mag}} = \sqrt{\left( {\frac{1}{n}{\sum\limits_{i = 1}^{n}I_{i}}} \right)^{2} + \left( {\frac{1}{n}{\sum\limits_{i = 1}^{n}Q_{i}}} \right)^{2}}} & {{EQN}\mspace{14mu} (5)}\end{matrix}$

In this equation, n is the number of samples of each symbol frequency, iis an index value, I_(i) represents the set of in phase magnitudes ofthe symbol frequencies and Q_(i) represents the set of quadraturemagnitudes of the symbol frequencies. Using this technique, Applicantshave recognized that the noise contributions measured as part of thein-phase and quadrature components tend to cancel one another. Incontrast, a prior art process is characterized as:

$\begin{matrix}{{{Symbol}\mspace{14mu} {Freq}\mspace{14mu} {Av}\mspace{14mu} {Mag}} = {\frac{1}{n}{\sum\limits_{i = 1}^{n}\sqrt{\left( {I_{i}^{2} + Q_{i}^{2}} \right)}}}} & {{EQN}\mspace{14mu} (6)}\end{matrix}$

Applicants submit that, because the I_(i) and Q_(i) values are subjectto noise, the average value that is produced by equation (6) tends torectify the noise and improperly add the noise to the resulting averagevalue. By using equation (5), the determined average values moreaccurately track the actual ambient noise and signal strength ofreceived symbols.

Based on FIGS. 28-31, as well as equation (5), it should be appreciatedthat Applicants have recognized and implemented an encode/decode processthat is extraordinarily robust and does not require the use of asynchronization symbol or symbols, or sync bits within the packetitself. By eliminating the need for sync symbols or bits as part of thepacket structure, the available bandwidth for data transmission isincreased. The process further exhibits noise immunity that is submittedto be heretofore unseen in the field of horizontal directional drillingand associated inground operations.

The foregoing description of the invention has been presented forpurposes of illustration and description. It is not intended to beexhaustive or to limit the invention to the precise form or formsdisclosed, and other modifications and variations may be possible inlight of the above teachings. Accordingly, those of skill in the artwill recognize certain modifications, permutations, additions andsub-combinations of the embodiments described above.

What is claimed is:
 1. A transmitter for use in conjunction with a horizontal directional drilling system that includes a drill string that extends from a drill rig to an inground tool that supports the transmitter such that extension and retraction of the drill string moves the inground tool through the ground during an inground operation, said transmitter comprising: an antenna; one or more sensors for generating one or more sensor signals; a processor configured for generating a multi-bit symbol stream based on the sensor signals; and an antenna driver arrangement for electrically driving the antenna based on the multi-bit symbol stream to emit an electromagnetic symbol stream at least for aboveground recovery of the sensor signals.
 2. The transmitter of claim 1 wherein the multi-bit symbol stream includes at least four different symbols such that each symbol represents at least two bits.
 3. The transmitter of claim 2 wherein each multi-bit symbol is transmitted at a symbol frequency that is unmodulated such that the different symbols are transmitted at different symbol frequencies.
 4. The transmitter of claim 3 wherein said processor is configured for transmitting each of the symbol frequencies at a specified power level based on a power allocation such that one of the symbol frequencies is transmitted at a first power level that is different than a second power level for at least one other symbol frequency.
 5. The transmitter of claim 1 further comprising: a direct digital synthesizer configured to generate the multi-bit symbol stream.
 6. The transmitter of claim 5 wherein the direct digital synthesizer is operable to generate a plurality of symbol frequencies across a wide frequency bandwidth and the direct digital synthesizer is configured to limit the symbol frequencies to a narrow bandwidth that is less than the wide frequency bandwidth and at least approximately matched to said antenna.
 7. The transmitter of claim 1 wherein the processor is configured for generating a depth signal and the antenna driver arrangement drives the antenna based on the depth signal to emit a dipole locating signal at least for use in determining a depth of the transmitter.
 8. The transmitter of claim 1 wherein the multi-bit symbol stream includes at least sixteen different symbols such that each symbol represents at least four bits.
 9. The transmitter of claim 1 wherein the processor is configured to transmit the multi-bit symbol stream subject to an error correction code for above ground reception of the multi-bit symbol stream by a receiver configured to determine a depth of the transmitter based at least in part on decoding the received multi-bit symbol stream subject to the error correction code.
 10. A portable device for use in conjunction with a transmitter that is configured to move through the ground in a region during an operational procedure while transmitting a transmitter signal that is receivable by the portable device subject to electromagnetic noise that can vary within said region, said portable device comprising: a receiver configured to receive the transmitter signal as a multibit symbol stream which at least characterizes a set of sensor information relating to the operation of the transmitter during the inground operation to recover the set of sensor information.
 11. The portable device of claim 10 wherein said receiver is configured to determine a depth of the transmitter during the inground operation based on reception of the multi-bit symbol frequencies.
 12. The portable device of claim 11 wherein the depth is determined based on a received amplitude of the multi-bit symbol frequencies.
 13. The portable device of claim 11 wherein the depth is determined based on multiplication of each multi-bit symbol frequency by an offset frequency.
 14. The portable device of claim 10 wherein the transmitter transmits the multi-bit symbol stream subject to an error correction code and the portable device determines the depth of the transmitter based at least in part on received amplitudes associated with corrected ones of the symbols in the received multi-bit symbol stream.
 15. A system for use in horizontal directional drilling that includes a drill string that extends from a drill rig to an inground tool such that extension and retraction of the drill string moves the inground tool through the ground during an inground operation, said system comprising: a transmitter that is supported by the inground tool including an antenna, one or more sensors for generating one or more sensor signals, a processor configured for generating a multi-bit symbol stream based on the sensor signals, and an antenna driver for electrically driving the antenna to emit an electromagnetic symbol stream based on the multi-bit symbol stream; and a portable device including a receiver configured to receive the multibit symbol stream in a normal mode during the inground operation to recover the set of sensor information subject to the electromagnetic noise.
 16. A system for use in horizontal directional drilling that includes a drill string that extends from a drill rig to an inground tool such that extension and retraction of the drill string moves the inground tool through the ground during an inground operation, said system comprising: a transmitter that includes one or more sensors for measuring one or more operational parameters characterizing the status of the inground tool, wherein such transmitter transmits at two or more frequencies with at least one of such frequencies itself representing multiple data bits characterizing said status of the inground tool irrespective of any modulation of such frequencies; and a portable device including a receiver configured to receive the two or more frequencies, and a processor configured to recover said status of the inground tool from the two or more frequencies.
 17. The system of claim 16 wherein said portable device is operable in a frequency selection mode to measure electromagnetic noise, absent the transmission of said frequencies, and identify frequencies based on the measured electromagnetic noise for subsequent transmission from said transmitter as said two or more frequencies.
 18. The system of claim 16 wherein each frequency represents two or more bits.
 19. The system of claim 18 wherein each frequency is unmodulated.
 20. A transmitter for use in horizontal directional drilling that includes a drill string that extends from a drill rig to an inground tool such that extension and retraction of the drill string moves the inground tool through the ground during an inground operation, said transmitter comprising: an antenna; one or more sensors for measuring one or more operational parameters characterizing the status of the inground tool; and an antenna driver for driving the antenna to transmit at two or more frequencies with at least one of such frequencies itself representing multiple data bits characterizing said status of the inground tool irrespective of any modulation of such frequencies for above ground receipt to recover the status of the inground tool.
 21. A portable device for use in conjunction with a transmitter that is configured to move through the ground in a region during an operational procedure while transmitting a transmitter signal that is receivable by the portable device subject to electromagnetic noise that can vary within said region, said portable device comprising: a receiver configured to receive two or more frequencies of the transmitter signal, at least one of which frequencies itself represents multiple data bits characterizing a status of the inground tool irrespective of any modulation; and a processor configured to recover the status of the inground tool from the two or more frequencies. 